inductance effect in acoustic design modeling
- By Pirotex2005
- Subwoofers
- 4 Replies
Hello. I have a question about modeling the acoustic design for a low-frequency speaker. I am confused by the different final frequency response with and without the "show effect of inductance" function. Which of these options will be closer to the real response?
Blue line with inductance effect. 2 dB differecnce at 55 Hz 😕
Blue line with inductance effect. 2 dB differecnce at 55 Hz 😕
Are Piezo Tweeters frowned upon?
I salvaged a pair of mini piezos and was thinking of adding them as super-tweeters.
Do they need a cap? They were originally installed behind 2.2uF?
Do they need a resistor? What does the resistor do?
Do they need a cap? They were originally installed behind 2.2uF?
Do they need a resistor? What does the resistor do?
ELEKIT TU-8888 Monoblock
ELEKIT TU-8888 Monoblock (available now)
Another wonderful products from Mr Fujita...PCB type DIY KIT...
Here below are the spec of the monoblock amp.
Tube usable : KT series from 88 to 170 (for output tubes) KT170 recommended
EF86 x 1 and ECC82 x 1 (driver tubes)
Max. output : Over 50W/unit (with KT170)
Manual bias adjustment - A meter at the top panel
No headphone feature
SP impedance : 4ohm and 8ohm
Input x 1
Balance adjustment volume (with 41-point click) on the rear side
Coupling capacitors : 3pcs (100n 450V x 1, 470n 630V x 2)
Max size for the coupling cap :
100n 450V ... D22 x L44 mm
470n 630V ... D20xL57 mm
* as of now and could change the spec a bit.
Tubes are not included.
*The amp is designed to use custom made Lundahl LL3733"
PCB for OPT and harnesses are included in the kit.
There are only 30 pairs or 60 pieces for USA/Canada in December
Price after Dec 15, 2024 $1,475.00 (will ship from Vancouver Canada)
Help needed on Regulated power supply on Class A 6L6 amp
- By mikeydee77
- Tubes / Valves
- 24 Replies
I would like some input on this.
I am repairing a build of this amp where it is humming badly as the regulator transistor is blown. The particular device is hard to obtain so I tried something similar npn of 120v rating and it worked for a while but then blew as well, I imagine due to over voltage as the B+ is 350v in operation and about 400v at switch on.
The build had worked ok for six months or so but I wonder if the design is a bit vulnerable and could be improved.
Are there any similar amp designs I should consider for inspiration on improving this?
Many thanks
LP storage stand build
- By DrewsBrews
- Everything Else
- 7 Replies
I've got a couple crates just sitting on the floor that I'm already outgrowing. So I needed something with a bit more capacity, and getting the vinyl up off the floor would be nice too...
I used the design here as inspiration. I don't anticipate needing that much capacity anytime soon, and I didn't want something that bulky sticking out into the room. Stacking single rows left-right simplified the design, omitting the center divider.
Lumber was sourced from Menards. I can give more details if needed.
The width of the lumber I used for the side panels and bottoms allowed for a small ledge out front to display the "on deck" sleeves. I ended up routing out a flute along the top edge to create a lip that keeps the sleeve from sliding onto the floor.
I decided to give the side panels some shape. I made a template/profile board and used a flushing bit on the router to transfer the shape to either side.
Simple dowel pin and pocket screw construction.
To prevent the bottom panels from sagging from the vinyl weight, and stiffen up the structure: I added some screws from underneath the bottom panels up into the front and back boards (not shown). It just wasn't needed for mockup purposes so I drilled the holes later.
I used the design here as inspiration. I don't anticipate needing that much capacity anytime soon, and I didn't want something that bulky sticking out into the room. Stacking single rows left-right simplified the design, omitting the center divider.
Lumber was sourced from Menards. I can give more details if needed.
The width of the lumber I used for the side panels and bottoms allowed for a small ledge out front to display the "on deck" sleeves. I ended up routing out a flute along the top edge to create a lip that keeps the sleeve from sliding onto the floor.
I decided to give the side panels some shape. I made a template/profile board and used a flushing bit on the router to transfer the shape to either side.
Simple dowel pin and pocket screw construction.
To prevent the bottom panels from sagging from the vinyl weight, and stiffen up the structure: I added some screws from underneath the bottom panels up into the front and back boards (not shown). It just wasn't needed for mockup purposes so I drilled the holes later.
-
Locked
An article that urges the continuing significance of the string-quartet form
- Music
- 1 Replies
The Synopsis:
"My thesis is that music written for the ensemble of two violins, one viola, and one cello constitutes the musical art form that most nearly approaches perfection, both in its past achievements and its present possibilities. The greatest minds and most perceptive dispositions find in the literature and performance of string-quartet music an endlessly self-renewing source of wonder and delight. The string quartet is a most felicitous meeting of potential and realization."
There are several YouTube embeds, and a link to a Qobuz playlist.
-
Locked
Philharmonic Audio BMR Monitor: "The Loudspeaker Bargain of the Century—At least so far!"
- Multi-Way
- 1 Replies
The pull quote above is the Subtitle of my Tracking Angle Review of the Philharmonic Audio BMR Monitor. BTW, the ArgentPur GaN-FET monobloc amps are AMAZING. Perhaps they are even "The Poor Man's darTZeel." That is, if Poor Men can afford $5,900/pr. SOTA power amps!!!
John Atkinson's measurements and Robert Schryer's review in Stereophile are required reading.
I have built more than a dozen loudspeakers of my own design (with help from outside experts!), and I can't grok (dating myself to the 1970s) how Philharmonic Audio can sell these for $2000/pair.
A cheap omni (aperiodic/weight on magnet)
- By cracked case
- Full Range
- 38 Replies
I noticed that Amazon UK had some In Phase Car Audio XTC17.2 woven glass fibre coned drivers for £ 21.11p, so I decided to order them and make some omnidirectional speakers with them
The driver is mounted on a small chamber that is connected to another, larger chamber by an aperiodic vent. My initial plan was to support the top chamber on coil springs with a foam seal, however the cast iron weights that became available ( the scrap bin at work ) are to heavy, and the temporary foam spacers that I used for initial testing work quite well
The driver is mounted on a small chamber that is connected to another, larger chamber by an aperiodic vent. My initial plan was to support the top chamber on coil springs with a foam seal, however the cast iron weights that became available ( the scrap bin at work ) are to heavy, and the temporary foam spacers that I used for initial testing work quite well
3D printing a large horn - ES-290 by Joseph Crowe
After finishing some electronics projects I'm now into speakers again. Call me crazy but I've really like to build something big this time. Currently my main system is build around a Limmer 630BC-flare with B&C DE360 and 6NDL38 - supported by a 10" Wavecor for the lows. Really nice whats coming out there - but you all know how life is going and we all are never finished with our projects.
So this time it will be something big and extravagant. I've bought the 3D drawing of the ES-290 by @Joseph Crowe yesterday and they are prepared for 3D printing already. Means: The big chonky horn is divided into eight pieces per half to get them on the bed of my printer. I'm using Fusion 360 for this. I've also prepared the single parts to be put together again using holes and dowels as you can see in the figure below.
It's looking neat and tiny in Fusion... But my printer bed is measuring 220x220 - know do your maths 😵. My CAD workstation got into some sweat during preparation.
Currently I'm doing some evaluation about printer settings, layer height and printing speed for the best quality with the least rework as possible (well the latter will always be a dream in reality - but let me stay dreaming for now). The evaluation models are printing using nearly no and the simplest infill for saving time (10% grid). It's just about how smooth it can get. The first try was using 0,5mm layer height with a 0,8mm nozzle (taking 14h for the bigger part in the picture below).
Not satisfying - so currently I'm doing another print using 0,3mm layers with a 0,6mm nozzle and slower speed - the example above was printed with 120mm/s. I'm at 60mm/s now and it's looking a lot more promising by now - but not as perfect as such an expensive and time consuming project could be in the end.
So I decided to use 0,16mm layers with a 0,6mm nozzle for doing the calculations for "final manufacturing" with a much bigger wall strength (set to 17x0,6mm -> 10,2mm walls) and stronger solid bottom- and top layers (10mm). I also calculated with much more and multidimensional infill - 50% gyroid for the direct exposed parts and equivalent less dense for the outer and rear parts should be enough.
Everything is prepared for using a single 1000g respectively 3000g roll filament per part as it can be seen above (green/yellow highlighted weight). I'll print using PLA filament because it's tought when using the correct settings and good for glueing without trouble. It's good behaving during post-processing (grinding, fill-ups), too. Last but not least it's easy to get in 3000g rolls too. PCTG would have been my favorite but it's not that easy to handle for glueing and there are not that many offers for 3000g rolls. I've been using PETG-CF in the past for adapters and baseplates in other projects but that filiament is not that good for doing smooth things like the inside of the horn.
So this time it will be something big and extravagant. I've bought the 3D drawing of the ES-290 by @Joseph Crowe yesterday and they are prepared for 3D printing already. Means: The big chonky horn is divided into eight pieces per half to get them on the bed of my printer. I'm using Fusion 360 for this. I've also prepared the single parts to be put together again using holes and dowels as you can see in the figure below.
It's looking neat and tiny in Fusion... But my printer bed is measuring 220x220 - know do your maths 😵. My CAD workstation got into some sweat during preparation.
Currently I'm doing some evaluation about printer settings, layer height and printing speed for the best quality with the least rework as possible (well the latter will always be a dream in reality - but let me stay dreaming for now). The evaluation models are printing using nearly no and the simplest infill for saving time (10% grid). It's just about how smooth it can get. The first try was using 0,5mm layer height with a 0,8mm nozzle (taking 14h for the bigger part in the picture below).
Not satisfying - so currently I'm doing another print using 0,3mm layers with a 0,6mm nozzle and slower speed - the example above was printed with 120mm/s. I'm at 60mm/s now and it's looking a lot more promising by now - but not as perfect as such an expensive and time consuming project could be in the end.
So I decided to use 0,16mm layers with a 0,6mm nozzle for doing the calculations for "final manufacturing" with a much bigger wall strength (set to 17x0,6mm -> 10,2mm walls) and stronger solid bottom- and top layers (10mm). I also calculated with much more and multidimensional infill - 50% gyroid for the direct exposed parts and equivalent less dense for the outer and rear parts should be enough.
Everything is prepared for using a single 1000g respectively 3000g roll filament per part as it can be seen above (green/yellow highlighted weight). I'll print using PLA filament because it's tought when using the correct settings and good for glueing without trouble. It's good behaving during post-processing (grinding, fill-ups), too. Last but not least it's easy to get in 3000g rolls too. PCTG would have been my favorite but it's not that easy to handle for glueing and there are not that many offers for 3000g rolls. I've been using PETG-CF in the past for adapters and baseplates in other projects but that filiament is not that good for doing smooth things like the inside of the horn.
Sony vFET Illustrated build guide
Nelson Pass testing procedure and instructions for DIY Sony Vfet amp -
Print this out!! 🙂
http://www.firstwatt.com/pdf/art_diy_sony_vfet.pdf
Pass DIY Sony Vfet amplifier illustrated build guide.
The Pass DIY Sony Vfet is a power amplifier from the brilliant mind of our technical, spiritual, and menu advisor, Nelson Pass. Quite possibly the most generous design yet, as Nelson released his private stash of the out of production 1970's Vfet to the DIY community so we can build an amp that nobody else can build nor buy. 🙂
How the circuit works
The Front End
The overall schematic of the VFET amp is somewhat complicated in spite of the simple gain path – this is due to the special DC regulation circuits to cater to the fussy VFET transistors in the output stage.
Fortunately the front end voltage gain stage of the amplifier is pretty simple.
The Vfet DIY Front-end is basically a mini-F5. It serves to give us enough voltage and current gain to drive the Vfets so we can wire the Vfet as followers (known as “Common Drain operation, where no voltage gain is acquired, only current).
The input signal starts at R1 (2,2K), which acts as a “gate-stopper” resistor for the input Jfets, as well as a current limiter in case the attached preamp has a big turn-on or -off thump. R2, (47K) is from the input of the Jfet to ground, this gives the Jfet gates a reference to ground, and also sets the input impedance, which is about 50K.
The signal goes through the Jfets (Q1,2) and the gain currents at their Drain outputs go through the cascode transistors (Q3,4) and then to drive the Gates of the voltage amplification (Vas) Mosfets (Q5,6). The cascode transistors, Q3 and Q4, act as a voltage umbrellas, protecting the Jfets from the voltages appearing at T11 and T13 so that their Drains experience a fraction of the DC and AC voltage T11 and T13 as determined by R7, R9 and R8, R10 driving Q3 and Q4.
To quote Mr. Pass, “Cascoding is a very useful technique where the gain transistor (in this case each of the Jfets) is coupled to another transistor operated in Common-Gate or Common-Base mode. This cascode transistor contributes very little of its own characteristic to the amplification, but it acts like a voltage umbrella, shielding the gain transistor from the high DC voltage and noise of the power supply rails.”
The Drain pins of Q5 and Q6 deliver the output voltage (but not enough current) to drive loudspeakers. This point is found at test node T18.
Mosfet Q5 and Q6 need some voltage applied to their gates in order to turn on (bias) and amplify signal, and this is provided by the voltage appearing across R35 + P3 and R34 +P4. P3 and P4 are there to adjust this voltage so as to set the right amount of current through Q4 and Q6. This right amount of current is defined by 1.5 V DC appearing across resistors R5 and R6 AND the output of the front end (T18) near 0 volts DC. So P3 and P4 not only adjust the idle current through the voltage gain transistors Q5 and Q6, but they also set the DC output that appears on their Drain pins.
Adjusting the Front End bias
Bias is set by measuring the DC voltage across R5, (test nodes T6-T7), and R6 (test nodes T8-T9). Proper bias is when you have 1.5V across each of these resistors, at the same time also having 0 volts DC offset, which is the voltage at T18 to GND.
Adjusting P3 and P4 will make changes in all three of those readings, so in my opinion, you must have three DC voltmeters to make this an easy job. Even the very inexpensive Chinese DMM will work fine for this, just make sure they have good, fresh batteries in them.
A single voltmeter will do that job, but it is more tedious. In any case you want to adjust each pot in small steps, going between the pots and voltage measurements in rotation.
You start with both potentiometers at full counterclockwise, which is the lowest bias setting. At first you are simply looking to get the voltages across R5 and R6 up to 1.5 in small steps. You turn P3 just a little bit, measuring a small change across R5, and then you do the same for P4 and R6. And then you go back to P3/R6 and repeat this process, raising the DC voltages slowly and in equal amounts.
You also want to check the DC voltage from T18 to ground as you do this – you want to keep it near 0V if possible.
I suggest you pause at around 1 V on R5 and R6 – as the Mosfets heat up, the current will rise on its own. Eventually you will find yourself with “warm” settings which give about 1.5V DC bias and close to 0 V DC output at T18.
How good is good enough? 1.3 V to 1.7V is OK for bias. +/-200 mV is good enough for offset at this point. We will tweak these values after the completed amplifier has run for a bit.
REMEMBER – BABY STEPS ON ALL ADJUSTMENTS.
If for some reason you can’t achieve full bias with zero offset, please troubleshoot in the build guide thread, and do tell us how much bias you can get with zero offset.
Please refer to the above schematic for component names
The output stage -
The Vfets themselves are wired as a complementary pair in source-follower mode. These add current gain to the signal sent to their Gate pins (to be able to drive speakers with a low output impedance) but have no voltage gain.
That’s it - the voltage gain has happened in the Front-End, and the Vfets just add current capability. 🙂 But it’s not quite that simple… is it?
Well, it actually is, until you realize that the DC powering up the Vfets must be regulated for a stable DC voltage and low noise, the same with the bias voltage. Also, both the rail voltage and the bias voltage have to power-up and power-down in a specific order to keep the Vfets from trying to draw a ton of current when the circuit is powered up.
What’s the proof of Nelson’s genius in this circuit? In how simple (and quite effective) the regulator sections have been designed. Let’s talk about that now.
Regulators
Each rail of each channel has 2 regulators, for a total of 4. Each of the two rails has a bias regulator and an output stage regulator. For clarity’s sake, I’m now going to talk about the positive rail’s regs - the negative rails have the same thing, just in a mirror image.
Both regulators use a TL431, a nice little 3-pin shunt reg that can have it’s output voltage set by 2 resistors. Looking at the schematic around Q11, you can see the unregulated voltage coming from the rail through a simple RC filter (R29 and C7) then through R13 (2.21K) and then the set resistors R11 (10K) and P1 (5K pot).
Because one of the two set resistors is a potentiometer wired as a variable resistor, changing that pot value changes the set voltage which provides Vfet voltage bias. It is very important that the bias voltage is place when power is applied to the Drain of the VFET, so this regulator circuit (bias reg) is designed to charge up much faster than the regulated voltage powering the Drain. (The output stage regulator)
Remember how it was mentioned earlier that the Sony Vfets are “Depletion” mode? They are normally on, letting current flow, and require a negative voltage applied to the gate to turn them off (this for the N type – The P channel devices require a positive voltage), or to otherwise set the value of the current (bias) through the VFET.
Looking at the regulated voltage at T19, (+14V) you want to note that it crosses over and connects to the gate of the negative P-channel Vfet Q10. The positive N-channel Vfet Q9, is likewise biased from the opposite rail. This is the opposite of what you are normally going to see with complementary Mosfet followers.
Moving on to the output regulators, you can see the circuitry near Q13 looks similar to the other reg, the exception being the voltage setting resistors are both fixed resistors. (R15, R17) These resistors and the TL431 make a voltage of 24V, seen at T2. This fixed voltage is controls the gate of a power Mosfet, Q7. This Mosfet is slaved to the regulator and lets the tiny TO-92 package regulator control many amps of current. Wonderful! This is necessary because the regulator needs to regulate the power going to the output stage which drives your speakers.
However there is no free lunch, and the Mosfet eats up about 4.2V (the Mosfet’s Vgs) in payment for doing it’s job. This is why you see a regulated rail voltage presented to the drains of the*Vfets of 19.8V (or so). An important part of the rail regulators is C1 (220uF), there to delay the arrival of this 19.8V to give the bias regulator time to ramp up. More about this here;
Power-up and power-down sequencing.
Back to depletion mode devices - The Vfets are normally on, right? We need to apply a bias voltage to control them. If the bias voltage isn’t there controlling current flow when the rail regulators power up, the Vfets will conduct an excessive amount of current and the 40-year old, completely irreplaceable smoke made from vintage Japanese un-obtainium, ground unicorn horn and pixie dust will escape. A sad result which will also smell bad.
So we must have the bias regulator working before the rail voltage is turned on. How did Nelson do it? A simple application of the time constant to charge 2 differently sized capacitors in the regulator circuits. The regulator with the smaller capacitor will turn on before the one with the larger cap. These caps in the race are C3 (47uF) on the bias reg and C1 (220uF) on the output regulator. It’s a 4.7-to-1 ratio and C3 will be charged about 4.7 times sooner.
The TL431 are shunt regulators, so they do not conduct current until the voltage rises to the regulation value, and then they conduct so as to hold that value.
So what about power down? The rail voltage needs to be turned off before the bias voltage. (the opposite of power-up) BUT, the big cap/small cap ratio will NOT work in our favor when shutting down. It would work exactly how you’d expect, where the small cap will discharge before the big one.
So what? The VFETs will drain the supply quickly enough, but when the supply voltage drops down quickly we may find the Gate to Source voltage (Vgs) of the Mosfet at a high enough voltage to stress the transistor into failure. It's not that likely, but everyone knows that Pass wears both a belt and suspenders to hold up his pants, and this is reflected in his design.
The fix is a simple 2-cent diode. On power down, D1 will be a very attractive and very low-resistance discharge path for C1, discharging it rapidly and turning Q7 (and Q13) off, and therefore the rail voltage to the Vfets. Because there is no special path for C3 in the bias regulator to discharge, it will operate for a longer time, which is perfectly ok.
The stuff in the middle.
Ok, so far everything has been explained except for the wiggles in the center of the schematic near T16 and T17. Remember how the bias voltage regulator is set by the divider made from R11 and P1? This is a DC value but we also need to let the AC have it’s say in the circuit, as the AC is our music. 🙂 The Front-End output signal was all big and powerful the last time we left it at T18. Now this signal needs to get to the Gates of the Vfets to move our speakers, and that signal (even though it’s AC) can get pretty darn big. Big enough that it’s voltage could effect the current flowing through the regulator set resistors. The network of C5, C6 and R21 through R24 prevent the music signal from having much influence on the bias regulators. Stable bias makes for happy Vfets, so this is a good thing.
What’s left… Um… R25 R26 (221ohm) are gatestoppers for the Vfet, R33 (22.1K) adjusts the brightness of the LED (which must be blue, this is a Pass amp, after all…) and R32 (0.1ohm) gives a place to measure the output stage current. (Bias)
Global feedback is applied from the output stage from the Vfets to the Jfets through R3 (2.21K), and then R4 (332ohm) to ground.
Cool! That's the basic overview of how it works. Let's build!!
😀 😀 😀
.
Print this out!! 🙂
http://www.firstwatt.com/pdf/art_diy_sony_vfet.pdf
Pass DIY Sony Vfet amplifier illustrated build guide.
The Pass DIY Sony Vfet is a power amplifier from the brilliant mind of our technical, spiritual, and menu advisor, Nelson Pass. Quite possibly the most generous design yet, as Nelson released his private stash of the out of production 1970's Vfet to the DIY community so we can build an amp that nobody else can build nor buy. 🙂
How the circuit works
The Front End
The overall schematic of the VFET amp is somewhat complicated in spite of the simple gain path – this is due to the special DC regulation circuits to cater to the fussy VFET transistors in the output stage.
Fortunately the front end voltage gain stage of the amplifier is pretty simple.
The Vfet DIY Front-end is basically a mini-F5. It serves to give us enough voltage and current gain to drive the Vfets so we can wire the Vfet as followers (known as “Common Drain operation, where no voltage gain is acquired, only current).
The input signal starts at R1 (2,2K), which acts as a “gate-stopper” resistor for the input Jfets, as well as a current limiter in case the attached preamp has a big turn-on or -off thump. R2, (47K) is from the input of the Jfet to ground, this gives the Jfet gates a reference to ground, and also sets the input impedance, which is about 50K.
The signal goes through the Jfets (Q1,2) and the gain currents at their Drain outputs go through the cascode transistors (Q3,4) and then to drive the Gates of the voltage amplification (Vas) Mosfets (Q5,6). The cascode transistors, Q3 and Q4, act as a voltage umbrellas, protecting the Jfets from the voltages appearing at T11 and T13 so that their Drains experience a fraction of the DC and AC voltage T11 and T13 as determined by R7, R9 and R8, R10 driving Q3 and Q4.
To quote Mr. Pass, “Cascoding is a very useful technique where the gain transistor (in this case each of the Jfets) is coupled to another transistor operated in Common-Gate or Common-Base mode. This cascode transistor contributes very little of its own characteristic to the amplification, but it acts like a voltage umbrella, shielding the gain transistor from the high DC voltage and noise of the power supply rails.”
The Drain pins of Q5 and Q6 deliver the output voltage (but not enough current) to drive loudspeakers. This point is found at test node T18.
Mosfet Q5 and Q6 need some voltage applied to their gates in order to turn on (bias) and amplify signal, and this is provided by the voltage appearing across R35 + P3 and R34 +P4. P3 and P4 are there to adjust this voltage so as to set the right amount of current through Q4 and Q6. This right amount of current is defined by 1.5 V DC appearing across resistors R5 and R6 AND the output of the front end (T18) near 0 volts DC. So P3 and P4 not only adjust the idle current through the voltage gain transistors Q5 and Q6, but they also set the DC output that appears on their Drain pins.
Adjusting the Front End bias
Bias is set by measuring the DC voltage across R5, (test nodes T6-T7), and R6 (test nodes T8-T9). Proper bias is when you have 1.5V across each of these resistors, at the same time also having 0 volts DC offset, which is the voltage at T18 to GND.
Adjusting P3 and P4 will make changes in all three of those readings, so in my opinion, you must have three DC voltmeters to make this an easy job. Even the very inexpensive Chinese DMM will work fine for this, just make sure they have good, fresh batteries in them.
A single voltmeter will do that job, but it is more tedious. In any case you want to adjust each pot in small steps, going between the pots and voltage measurements in rotation.
You start with both potentiometers at full counterclockwise, which is the lowest bias setting. At first you are simply looking to get the voltages across R5 and R6 up to 1.5 in small steps. You turn P3 just a little bit, measuring a small change across R5, and then you do the same for P4 and R6. And then you go back to P3/R6 and repeat this process, raising the DC voltages slowly and in equal amounts.
You also want to check the DC voltage from T18 to ground as you do this – you want to keep it near 0V if possible.
I suggest you pause at around 1 V on R5 and R6 – as the Mosfets heat up, the current will rise on its own. Eventually you will find yourself with “warm” settings which give about 1.5V DC bias and close to 0 V DC output at T18.
How good is good enough? 1.3 V to 1.7V is OK for bias. +/-200 mV is good enough for offset at this point. We will tweak these values after the completed amplifier has run for a bit.
REMEMBER – BABY STEPS ON ALL ADJUSTMENTS.
If for some reason you can’t achieve full bias with zero offset, please troubleshoot in the build guide thread, and do tell us how much bias you can get with zero offset.
Please refer to the above schematic for component names
The output stage -
The Vfets themselves are wired as a complementary pair in source-follower mode. These add current gain to the signal sent to their Gate pins (to be able to drive speakers with a low output impedance) but have no voltage gain.
That’s it - the voltage gain has happened in the Front-End, and the Vfets just add current capability. 🙂 But it’s not quite that simple… is it?
Well, it actually is, until you realize that the DC powering up the Vfets must be regulated for a stable DC voltage and low noise, the same with the bias voltage. Also, both the rail voltage and the bias voltage have to power-up and power-down in a specific order to keep the Vfets from trying to draw a ton of current when the circuit is powered up.
What’s the proof of Nelson’s genius in this circuit? In how simple (and quite effective) the regulator sections have been designed. Let’s talk about that now.
Regulators
Each rail of each channel has 2 regulators, for a total of 4. Each of the two rails has a bias regulator and an output stage regulator. For clarity’s sake, I’m now going to talk about the positive rail’s regs - the negative rails have the same thing, just in a mirror image.
Both regulators use a TL431, a nice little 3-pin shunt reg that can have it’s output voltage set by 2 resistors. Looking at the schematic around Q11, you can see the unregulated voltage coming from the rail through a simple RC filter (R29 and C7) then through R13 (2.21K) and then the set resistors R11 (10K) and P1 (5K pot).
Because one of the two set resistors is a potentiometer wired as a variable resistor, changing that pot value changes the set voltage which provides Vfet voltage bias. It is very important that the bias voltage is place when power is applied to the Drain of the VFET, so this regulator circuit (bias reg) is designed to charge up much faster than the regulated voltage powering the Drain. (The output stage regulator)
Remember how it was mentioned earlier that the Sony Vfets are “Depletion” mode? They are normally on, letting current flow, and require a negative voltage applied to the gate to turn them off (this for the N type – The P channel devices require a positive voltage), or to otherwise set the value of the current (bias) through the VFET.
Looking at the regulated voltage at T19, (+14V) you want to note that it crosses over and connects to the gate of the negative P-channel Vfet Q10. The positive N-channel Vfet Q9, is likewise biased from the opposite rail. This is the opposite of what you are normally going to see with complementary Mosfet followers.
Moving on to the output regulators, you can see the circuitry near Q13 looks similar to the other reg, the exception being the voltage setting resistors are both fixed resistors. (R15, R17) These resistors and the TL431 make a voltage of 24V, seen at T2. This fixed voltage is controls the gate of a power Mosfet, Q7. This Mosfet is slaved to the regulator and lets the tiny TO-92 package regulator control many amps of current. Wonderful! This is necessary because the regulator needs to regulate the power going to the output stage which drives your speakers.
However there is no free lunch, and the Mosfet eats up about 4.2V (the Mosfet’s Vgs) in payment for doing it’s job. This is why you see a regulated rail voltage presented to the drains of the*Vfets of 19.8V (or so). An important part of the rail regulators is C1 (220uF), there to delay the arrival of this 19.8V to give the bias regulator time to ramp up. More about this here;
Power-up and power-down sequencing.
Back to depletion mode devices - The Vfets are normally on, right? We need to apply a bias voltage to control them. If the bias voltage isn’t there controlling current flow when the rail regulators power up, the Vfets will conduct an excessive amount of current and the 40-year old, completely irreplaceable smoke made from vintage Japanese un-obtainium, ground unicorn horn and pixie dust will escape. A sad result which will also smell bad.
So we must have the bias regulator working before the rail voltage is turned on. How did Nelson do it? A simple application of the time constant to charge 2 differently sized capacitors in the regulator circuits. The regulator with the smaller capacitor will turn on before the one with the larger cap. These caps in the race are C3 (47uF) on the bias reg and C1 (220uF) on the output regulator. It’s a 4.7-to-1 ratio and C3 will be charged about 4.7 times sooner.
The TL431 are shunt regulators, so they do not conduct current until the voltage rises to the regulation value, and then they conduct so as to hold that value.
So what about power down? The rail voltage needs to be turned off before the bias voltage. (the opposite of power-up) BUT, the big cap/small cap ratio will NOT work in our favor when shutting down. It would work exactly how you’d expect, where the small cap will discharge before the big one.
So what? The VFETs will drain the supply quickly enough, but when the supply voltage drops down quickly we may find the Gate to Source voltage (Vgs) of the Mosfet at a high enough voltage to stress the transistor into failure. It's not that likely, but everyone knows that Pass wears both a belt and suspenders to hold up his pants, and this is reflected in his design.
The fix is a simple 2-cent diode. On power down, D1 will be a very attractive and very low-resistance discharge path for C1, discharging it rapidly and turning Q7 (and Q13) off, and therefore the rail voltage to the Vfets. Because there is no special path for C3 in the bias regulator to discharge, it will operate for a longer time, which is perfectly ok.
The stuff in the middle.
Ok, so far everything has been explained except for the wiggles in the center of the schematic near T16 and T17. Remember how the bias voltage regulator is set by the divider made from R11 and P1? This is a DC value but we also need to let the AC have it’s say in the circuit, as the AC is our music. 🙂 The Front-End output signal was all big and powerful the last time we left it at T18. Now this signal needs to get to the Gates of the Vfets to move our speakers, and that signal (even though it’s AC) can get pretty darn big. Big enough that it’s voltage could effect the current flowing through the regulator set resistors. The network of C5, C6 and R21 through R24 prevent the music signal from having much influence on the bias regulators. Stable bias makes for happy Vfets, so this is a good thing.
What’s left… Um… R25 R26 (221ohm) are gatestoppers for the Vfet, R33 (22.1K) adjusts the brightness of the LED (which must be blue, this is a Pass amp, after all…) and R32 (0.1ohm) gives a place to measure the output stage current. (Bias)
Global feedback is applied from the output stage from the Vfets to the Jfets through R3 (2.21K), and then R4 (332ohm) to ground.
Cool! That's the basic overview of how it works. Let's build!!
😀 😀 😀
.
Discrete opamp with zero input bias current
- By lineup
- Solid State
- 3 Replies
It is possible to compensate so the input current becomes zero.
I don't really know what this should be good for.
At least there is one capacitor less 😀
This circuit simulates THD 0.000062% at 1Vrms into 1kOhm.
I don't really know what this should be good for.
At least there is one capacitor less 😀
This circuit simulates THD 0.000062% at 1Vrms into 1kOhm.
Sansui AU-D9 – Power transistors not amplifying after restoration
- By GESMAN
- Solid State
- 44 Replies
Attachments
ADSP21489 + 2in / 8out PCM1798 board
- By khoidang1107
- Introductions
- 1 Replies
Hello everyone, can anyone help me with the DP48X driver and FIR48 firmware? I also don't know if my board is version 1.0 or 2.0. I hope everyone can help me. Thanks for reading.
DIY stiffening cone techniques - How to
- By Freedom666
- Construction Tips
- 38 Replies
How to DIY making a paper cone (or others) more stiff.
Here first example:
fast drying hairspray liquid
Never apply too much at once so the paper fibers do not become wet and weak and throw wrinkles!
Better less but several times applying
then if possible fast dry in the sun
Here first example:
fast drying hairspray liquid
Never apply too much at once so the paper fibers do not become wet and weak and throw wrinkles!
Better less but several times applying
then if possible fast dry in the sun
Design Phase - 35Hz Tapped Horn w/ Compression Chamber (Hog Scoop + Inverted Driver)
- By Databass
- Subwoofers
- 3 Replies
Design Phase - 35Hz Tapped Horn w/ Compression Chamber (Hog Scoop + Inverted Driver)
This project has been inspired by all the love for Hog Scoops... And the fair share of hate they get as well.
This will bee my first attempt at an 18" horn subwoofer.
Proposed plan is based on the well known "Hog Scoop",
a tried and tested free design hosted over on the freespeakerplans.com forum.
https://www.freespeakerplans.com/?view=article&id=31:hog-scoop
Drawn up nearly 20 years ago, this well proven classic has stood the test of time and is a very
clever piece of speaker box design, still to this day.
Credit for this creation must go to Stipe Ercegovic, more commonly known a Staiper, a well respected member on the Speakerplans Scoop subforum for a long time,
but he disappearred from there several years ago and doesn't seem to frequent these places anymore.
The Hog Scoop, technically an F1 hybrid of sorts, the offspring of two genetically similar species,
living a life somewhere between a tapped horn and a traditional back loaded "scoop".
A rear loaded horn with the bonus of the driver shooting into the horn path, or a tapped horn enclosure
equipped with a compression chamber.... Call it what you will, this quarter wave resonator is well capable of
hitting the low bass notes and producing high SPL when equipped with the right hardware.
So, I would like to run inverted drivers in a slightly tweaked version of this enclosure, something similar to the image attached to the end of this post.
These are the M4.18 subwoofers, made by MM-Acoustics in Europe, a Macedonian outfit headed by Marjan Milosevic, a regular poster and long time member over on SP.
My god, they look absolutely amazing, so business like.
This is the look I am going for.... They are absolutely badass indeed.
Anyway..... enough airy introductions, and down to business then !!
Hog Scoop Hornresp input parameters
First things first, I decided it would bee prudent to simulate a hog scoop with a driver in it's normal orientation first..... Cone side out.
This was to bee my baseline sim, a benchmark to work from before l went tweaking the original design.
And staight away I ran into problems.
For a start, the speakerplans.com forum isn't very active these days..... And a huge amount of information from back when it was is just lost.
Many externally hosted images are not there any more, there are many dead links, hosted plans, you name it, gone !!!
It didn't make my research easy but I did what I could with what I had.
That said, there were many helpful and informative threads to bee found amongst all the bickericking and the politics in the Scoop subforum.
So, lets discuss what was available to find on the net first.... Multiple posts regarding the Hog Scoop hornresp input parameters were scattered around the web....
And just not consistent.
I was hoping to pull some numbers out of a post or a sketch somewhere but straight off the bat, numbers and posts were conflicting.
Unfortunately, I found more than one set of digits, which only infuriated and confused me even more.
Here are the most posted (and reposted) HR Hog Scoop numbers, I beleive they orignally came from Staiper, and these ones were the most common to bee found online,
copy and pasted in at least 4 different places, so I am inclined to stick with the guy who originally penned the Hog Scoop design. He did create it after all.
Stipe Hog Scoop
S1 = 750
Con = 112.8
S2 = 1062
Con = 42
S3 = 1770
Con = 51
S4 = 2544
Con = 47
S5 = 5700
Vrc = 0
Lrc = 0
Fr = 0
Tal = 0
Vtc = 49456
Atc = 86
Re-posted by bitzo,
https://forum.speakerplans.com/forum_posts.asp?TID=16974&PID=214119&title=database-simulation#214119
toastyghost,
https://forum.speakerplans.com/forum_posts.asp?TID=16974&PID=727583&title=database-simulation#727583
Bee,
https://forum.speakerplans.com/forum_posts.asp?TID=72607&PID=828911&title=hog-scoop-drivers#828911
and salzburgsound system
The next set of input data numbers to bee found were these ones you see below.....
--------->
S1 = 750
Par = 0.10
S2 = 750
Par = 200.90
S3 = 1700
Par = 61.00
S4 = 4740
Par = 12.10
S5 = 5700
Vrc = 0
Ap1 = 0
Lrc = 0
Lpt = 0
Vtc = 37060
Atc = 1453.10
Posted by Heathrow B Line
https://forum.speakerplans.com/foru...=901195&title=scoop-hr-reference-guide#901195
epa
Freddi
And one more, with newer and revised numbers, taken from a post by user "epa" from 2014.
S1 = 750
Par = 0.10
S2 = 750
Par = 187.90
S3 = 1700
Par = 52.00
S4 = 4740
Par = 12.10
S5 = 5700
Vrc = 0
Lrc = 0
Ap1 = 0
Lpt = 0
Vtc = 41091
Atc = 2210.00
Posted by epa
Other than these three sets of input parameters, I did find a few posts by Staiper himself, modeling the Hog Scoop as an exponential horn,
in quite a numerically sparce fashion I must add.... It would seem this stripped down form of modelling came from Rog Mogale himself.
i will link after the following
---------->
Staiper Exponential Hog
S1 = 750
S2 = 5700
L = 260 EXP
Vtc = 36000 (with driver)
Atc = 1000
https://forum.speakerplans.com/forum_posts.asp?TID=8887&PID=218876&title=hog-scoop#218876
Staiper Exponential Again....
S1 = 750
S2 = 5700
L = 257 EXP
Vtc = 37000
Atc = 9000
https://forum.speakerplans.com/forum_posts.asp?TID=8329&PID=77534&title=tapped-horn-design#77534
And finally....... An attempt at simming a Hog Scoop by the man, Rog Mogale himself.
S1 = 750
S2 = 5700
L = 240 EXP
Vtc = 35000
Atc = 890
https://forum.speakerplans.com/forum_posts.asp?TID=8329&PID=77097&title=tapped-horn-design#77097
And here is a good thread full of interesting tit bits where Rog discusses his exponential scoop modelling methods with others.
https://forum.speakerplans.com/forum_posts.asp?TID=8329&PID=77097&title=tapped-horn-design#77097
A must read for anyone into this line of work.
Annd..... Time to take stock.
All my research and reading paid off, but unfortunately I still feel I do not have any concrete numbers for a Hog Scoop simulation binge.
So, I took some time out and drew up a scaled plan of the Hog, just enough to play with.
it gave me a much more definite idea of the direction I asm going in, speaker boxes ofthen quite literally build themselves as the project goes on.
Solutions are often just so obvious when you have the material object taking shape before your eyes.
And measurements can bee taken from the plan and scaled up to something that resembles real world numbers.
(Drawing attached)
And just to bee sure, I am going to get my hands on one side panel first, (18mm sheet board 986 x 950mm) and draw it out @ a life size scale on the floor.
Drivers for this design
I currently have 2 B&C 18TBX100 (4 ohm) belonging to a friend to use for the first 2 boxes and am debating which drivers to buy for the rest.
Quite likely B&C as well, I like the power handling (1500w) and price of the 18SW1OO.
When given the option, I usually buy 4 ohm versions of any driver when I can.
They can often bee had for the same price as their 8ohm equivalents but sometimes not. Sometimes they are just more hassle to obtain but I do make an effort to
use 4 ohm drivers when ever I can.
I am also open to other brands, I do know I need a certain type of driver to do well in these boxes and if one particular driver turns out to bee
especially suited to the task it will surely bee strongly considered.
So far, everything is going well.... The only other uncertainty I have right now is the design of the rear chamber.
I am not sure exactly how it will end up but I am confident the design (and build) process will answer it's own questions if and when they arise.
More on that in detail later.
Okay, I think I have written enough for now.... Any advice and criticism is warmly welcomed.
Please feel free to join me on my journey of discovery and don't bee shy to interject or comment if need bee.
This project has been inspired by all the love for Hog Scoops... And the fair share of hate they get as well.
This will bee my first attempt at an 18" horn subwoofer.
Proposed plan is based on the well known "Hog Scoop",
a tried and tested free design hosted over on the freespeakerplans.com forum.
https://www.freespeakerplans.com/?view=article&id=31:hog-scoop
Drawn up nearly 20 years ago, this well proven classic has stood the test of time and is a very
clever piece of speaker box design, still to this day.
Credit for this creation must go to Stipe Ercegovic, more commonly known a Staiper, a well respected member on the Speakerplans Scoop subforum for a long time,
but he disappearred from there several years ago and doesn't seem to frequent these places anymore.
The Hog Scoop, technically an F1 hybrid of sorts, the offspring of two genetically similar species,
living a life somewhere between a tapped horn and a traditional back loaded "scoop".
A rear loaded horn with the bonus of the driver shooting into the horn path, or a tapped horn enclosure
equipped with a compression chamber.... Call it what you will, this quarter wave resonator is well capable of
hitting the low bass notes and producing high SPL when equipped with the right hardware.
So, I would like to run inverted drivers in a slightly tweaked version of this enclosure, something similar to the image attached to the end of this post.
These are the M4.18 subwoofers, made by MM-Acoustics in Europe, a Macedonian outfit headed by Marjan Milosevic, a regular poster and long time member over on SP.
My god, they look absolutely amazing, so business like.
This is the look I am going for.... They are absolutely badass indeed.
Anyway..... enough airy introductions, and down to business then !!
Hog Scoop Hornresp input parameters
First things first, I decided it would bee prudent to simulate a hog scoop with a driver in it's normal orientation first..... Cone side out.
This was to bee my baseline sim, a benchmark to work from before l went tweaking the original design.
And staight away I ran into problems.
For a start, the speakerplans.com forum isn't very active these days..... And a huge amount of information from back when it was is just lost.
Many externally hosted images are not there any more, there are many dead links, hosted plans, you name it, gone !!!
It didn't make my research easy but I did what I could with what I had.
That said, there were many helpful and informative threads to bee found amongst all the bickericking and the politics in the Scoop subforum.
So, lets discuss what was available to find on the net first.... Multiple posts regarding the Hog Scoop hornresp input parameters were scattered around the web....
And just not consistent.
I was hoping to pull some numbers out of a post or a sketch somewhere but straight off the bat, numbers and posts were conflicting.
Unfortunately, I found more than one set of digits, which only infuriated and confused me even more.
Here are the most posted (and reposted) HR Hog Scoop numbers, I beleive they orignally came from Staiper, and these ones were the most common to bee found online,
copy and pasted in at least 4 different places, so I am inclined to stick with the guy who originally penned the Hog Scoop design. He did create it after all.
Stipe Hog Scoop
S1 = 750
Con = 112.8
S2 = 1062
Con = 42
S3 = 1770
Con = 51
S4 = 2544
Con = 47
S5 = 5700
Vrc = 0
Lrc = 0
Fr = 0
Tal = 0
Vtc = 49456
Atc = 86
Re-posted by bitzo,
https://forum.speakerplans.com/forum_posts.asp?TID=16974&PID=214119&title=database-simulation#214119
toastyghost,
https://forum.speakerplans.com/forum_posts.asp?TID=16974&PID=727583&title=database-simulation#727583
Bee,
https://forum.speakerplans.com/forum_posts.asp?TID=72607&PID=828911&title=hog-scoop-drivers#828911
and salzburgsound system
The next set of input data numbers to bee found were these ones you see below.....
--------->
S1 = 750
Par = 0.10
S2 = 750
Par = 200.90
S3 = 1700
Par = 61.00
S4 = 4740
Par = 12.10
S5 = 5700
Vrc = 0
Ap1 = 0
Lrc = 0
Lpt = 0
Vtc = 37060
Atc = 1453.10
Posted by Heathrow B Line
https://forum.speakerplans.com/foru...=901195&title=scoop-hr-reference-guide#901195
epa
Freddi
And one more, with newer and revised numbers, taken from a post by user "epa" from 2014.
S1 = 750
Par = 0.10
S2 = 750
Par = 187.90
S3 = 1700
Par = 52.00
S4 = 4740
Par = 12.10
S5 = 5700
Vrc = 0
Lrc = 0
Ap1 = 0
Lpt = 0
Vtc = 41091
Atc = 2210.00
Posted by epa
Other than these three sets of input parameters, I did find a few posts by Staiper himself, modeling the Hog Scoop as an exponential horn,
in quite a numerically sparce fashion I must add.... It would seem this stripped down form of modelling came from Rog Mogale himself.
i will link after the following
---------->
Staiper Exponential Hog
S1 = 750
S2 = 5700
L = 260 EXP
Vtc = 36000 (with driver)
Atc = 1000
https://forum.speakerplans.com/forum_posts.asp?TID=8887&PID=218876&title=hog-scoop#218876
Staiper Exponential Again....
S1 = 750
S2 = 5700
L = 257 EXP
Vtc = 37000
Atc = 9000
https://forum.speakerplans.com/forum_posts.asp?TID=8329&PID=77534&title=tapped-horn-design#77534
And finally....... An attempt at simming a Hog Scoop by the man, Rog Mogale himself.
S1 = 750
S2 = 5700
L = 240 EXP
Vtc = 35000
Atc = 890
https://forum.speakerplans.com/forum_posts.asp?TID=8329&PID=77097&title=tapped-horn-design#77097
And here is a good thread full of interesting tit bits where Rog discusses his exponential scoop modelling methods with others.
https://forum.speakerplans.com/forum_posts.asp?TID=8329&PID=77097&title=tapped-horn-design#77097
A must read for anyone into this line of work.
Annd..... Time to take stock.
All my research and reading paid off, but unfortunately I still feel I do not have any concrete numbers for a Hog Scoop simulation binge.
So, I took some time out and drew up a scaled plan of the Hog, just enough to play with.
it gave me a much more definite idea of the direction I asm going in, speaker boxes ofthen quite literally build themselves as the project goes on.
Solutions are often just so obvious when you have the material object taking shape before your eyes.
And measurements can bee taken from the plan and scaled up to something that resembles real world numbers.
(Drawing attached)
And just to bee sure, I am going to get my hands on one side panel first, (18mm sheet board 986 x 950mm) and draw it out @ a life size scale on the floor.
Drivers for this design
I currently have 2 B&C 18TBX100 (4 ohm) belonging to a friend to use for the first 2 boxes and am debating which drivers to buy for the rest.
Quite likely B&C as well, I like the power handling (1500w) and price of the 18SW1OO.
When given the option, I usually buy 4 ohm versions of any driver when I can.
They can often bee had for the same price as their 8ohm equivalents but sometimes not. Sometimes they are just more hassle to obtain but I do make an effort to
use 4 ohm drivers when ever I can.
I am also open to other brands, I do know I need a certain type of driver to do well in these boxes and if one particular driver turns out to bee
especially suited to the task it will surely bee strongly considered.
So far, everything is going well.... The only other uncertainty I have right now is the design of the rear chamber.
I am not sure exactly how it will end up but I am confident the design (and build) process will answer it's own questions if and when they arise.
More on that in detail later.
Okay, I think I have written enough for now.... Any advice and criticism is warmly welcomed.
Please feel free to join me on my journey of discovery and don't bee shy to interject or comment if need bee.
Building stands for a pair of Monitor Audio r352 speakers
- By B O A B
- Construction Tips
- 3 Replies
Hello,
I recently picked up a pair of Monitor Audio r352 speakers of eBay, but they didn't come with any stands. I like the look of the stands that came with these back in the day (see the attached photo), but I can't find any information about them online. I reached out to a couple of eBay sellers who were selling the speakers with stands, but their either didn't reply, or couldn't help. I even reached out to Monitor Audio, who after a while got back to me to say they also had no information in the archive, nor did they have a pair in storage.
It's a long shot I know, but I don't suppose anyone on here has a pair of the stands and could supply me with the relevant measurements...?
If not, can anyone recommend some rules of thumb, around the size and thickness of both the base plate and column? The base of the speaker cabinets have captive nuts, so I guess the top plate size if pretty much self explanatory. The only other thing I really need is some for some guidance over the angle the speaker sits at.
Any construction tips for stands? Is it simply a case of screwing the column to both plates and calling it done? Or is there a better way of joining everything to help dissipate, or mitigate, vibrations coming up from the floor?
Any help would be much appreciated.
I recently picked up a pair of Monitor Audio r352 speakers of eBay, but they didn't come with any stands. I like the look of the stands that came with these back in the day (see the attached photo), but I can't find any information about them online. I reached out to a couple of eBay sellers who were selling the speakers with stands, but their either didn't reply, or couldn't help. I even reached out to Monitor Audio, who after a while got back to me to say they also had no information in the archive, nor did they have a pair in storage.
It's a long shot I know, but I don't suppose anyone on here has a pair of the stands and could supply me with the relevant measurements...?
If not, can anyone recommend some rules of thumb, around the size and thickness of both the base plate and column? The base of the speaker cabinets have captive nuts, so I guess the top plate size if pretty much self explanatory. The only other thing I really need is some for some guidance over the angle the speaker sits at.
Any construction tips for stands? Is it simply a case of screwing the column to both plates and calling it done? Or is there a better way of joining everything to help dissipate, or mitigate, vibrations coming up from the floor?
Any help would be much appreciated.
ICEpower 1000A help to identify parts missing
- By hunter audio
- Class D
- 13 Replies
Hi All
Please help to identify the parts marked in the Ice Power 1000a module picture they are adjacent to the input connector. I have 2 modules one the parts have unfortunately come off and got lost in transit and one module with the parts populated. Most of the components are marked and identifiable, somehow the parts that went missing are not marked.
Below is the module with the parts missing in blue
Below is a picture of the module that is with all the components.
Thanks.
Please help to identify the parts marked in the Ice Power 1000a module picture they are adjacent to the input connector. I have 2 modules one the parts have unfortunately come off and got lost in transit and one module with the parts populated. Most of the components are marked and identifiable, somehow the parts that went missing are not marked.
Below is the module with the parts missing in blue
Below is a picture of the module that is with all the components.
Thanks.
"Pangolins", a Dayton 10" 2.5way...
- By wolf_teeth
- Multi-Way
- 26 Replies
Attachments
-
PangolinsFinalC0deg.png -
PangolinsFinalC0degHD.png -
PangolinsFinalCOffAxes0-60.png -
PangolinBox.png -
PangolinsBoxDrawing.png -
PangolinXmax.png -
Dayton RST28F-4_1Mod.jpg -
Dayton RST28F-4_2Mod.jpg -
Mod1HD.jpg -
Mod2HD.jpg -
RSTcup.png -
2ndtuningdiscCab1withCoil.png -
2ndtuningdiscCab2withCoil.png -
Pangolin_2pt5-2pt0_8pt2-52ohm-0.7-notch_OnTweeterAxis_halfmeter.png -
PangolinCab1-RST28FMod-OnAxis.png -
PangolinCab1-SD270A88-OnTweeterAxis.png -
PangolinCab1-SD270A88-OnTweeterAxisHD.png -
PangolinCab2-SD270A88-OnTweeterAxisHD.png -
fgb0td6ca5sd.jpg -
Pangolins_D_AbsolutelyFinalXover.png
Current feedback Mosfet Circlotron
- By UltimateX86
- Solid State
- 267 Replies
Current feedback Mosfet Circlotron : it's alive !
Hi,
I present you the amp that I just realized.
This is a current feedback Mosfet Circlotron
Without Diyaudio forum and another French, this amp could never be created.
So thank you all the people who helped me and especially Joe Berry who gave me the Keystone of this amp, the current generator so particular that guarantee a thermal stability
A special thanks to gionag, my friend Rabbi Michele, who is the first to have made this amp and who made this great layout
YouTube : low cost phone record... but it's alive !
in real R15=500, R14=220, C3-C4= 47p
YouTube : low cost phone record... but it's alive !
Hi,
I present you the amp that I just realized.
This is a current feedback Mosfet Circlotron
Without Diyaudio forum and another French, this amp could never be created.
So thank you all the people who helped me and especially Joe Berry who gave me the Keystone of this amp, the current generator so particular that guarantee a thermal stability A special thanks to gionag, my friend Rabbi Michele, who is the first to have made this amp and who made this great layout YouTube : low cost phone record... but it's alive !
in real R15=500, R14=220, C3-C4= 47p
YouTube : low cost phone record... but it's alive !
For Sale GZLOZone preamp
For sale is a GZLOZone PRT-01A populated preamp board with tubes and power transformer for sale. Just add RCA jacks and volume pot. I don't have any technical info as this was given to me by a friend. I will ship to you for $30 including shipping, most likely USPS priority mail flat rate box, CONUS only.
Open source Waveguides for CNC & 3D printing!
I've been working on a waveguide project where I would develop high performance waveguides for some popular tweeters, and than share the files for anyone to print or CNC as they are able. The discussion started here: 3D printed waveguides - Techtalk Speaker Building, Audio, Video Discussion Forum for more details. The short summary is starting with the SB26ADC and a 5" waveguide, I varied flare radius, phase shields, depths, etc. I've started to find some best practices (at least with this dome/horn combo) and all future guides will have elliptical mouths, and with and without phase shields. I then skipped to 8" designs to verify those best practices scale, so anything in between should be trivial to design and still have the expected performance. Tweeters that I'm currently designing for are: SBA SB26ADC, SBA SB19ST, Satori TW29RN, and Transducer Labs N26C-A.
To get some frame of reference, here measurements of the Kef Reference 5 prototype, and Revel Concerta2 (which I suspect also uses the SB26ADC):
Here is my "C" waveguide with SB26ADC, with and without phase shield:
Quite good results I think, better than Revel achieved with the same tweeter! Anyway, the reason I am posting now though this project is still in its infancy, is I've connected with a couple gentleman with the ability to model waveguides. This could speed things up considerably, as waveguide design is extremely iterative otherwise because 90% of the performance happens at the dome/throat interface and this is not easily modeled.
We've started with verifying that the sim can accurately predict reality. With as close a representation as possible of the dome here are the results:
Extremely encouraging! I'd like to do some additional verification - particularly of the 8" guides I've started working on - and then I hope to push the sims to the max, and start producing some really excellent guides for popular tweeters.
To get some frame of reference, here measurements of the Kef Reference 5 prototype, and Revel Concerta2 (which I suspect also uses the SB26ADC):
Here is my "C" waveguide with SB26ADC, with and without phase shield:
Quite good results I think, better than Revel achieved with the same tweeter! Anyway, the reason I am posting now though this project is still in its infancy, is I've connected with a couple gentleman with the ability to model waveguides. This could speed things up considerably, as waveguide design is extremely iterative otherwise because 90% of the performance happens at the dome/throat interface and this is not easily modeled.
We've started with verifying that the sim can accurately predict reality. With as close a representation as possible of the dome here are the results:
Extremely encouraging! I'd like to do some additional verification - particularly of the 8" guides I've started working on - and then I hope to push the sims to the max, and start producing some really excellent guides for popular tweeters.
modeling a loudspeaker using transfer matrices
- By parallelpen
- Software Tools
- 12 Replies
Hi everyone,
I am trying to make my own simulating tool using GNU Octave and transfer matrices methode. I am encountering difficulties in validating my simple model (infinite baffle) because it does not match hornresp results. i think i am not that far from a correct model but not here yet. as you can see phase and SPL are not the same as hornresp.
below is the code with some comments. Note that i do not think problem come from my radiatoin impedance since i used this function for analytical modeling and results mached hornresp with my analytical model.
I am trying to make my own simulating tool using GNU Octave and transfer matrices methode. I am encountering difficulties in validating my simple model (infinite baffle) because it does not match hornresp results. i think i am not that far from a correct model but not here yet. as you can see phase and SPL are not the same as hornresp.
below is the code with some comments. Note that i do not think problem come from my radiatoin impedance since i used this function for analytical modeling and results mached hornresp with my analytical model.
Code:
clear all; close all; clc;
[c, rho, P, P_ref, Air_Imp, f, w] = Generals(); % initialise constantes and frequenies
wc = w / c; % wave number
driver = 'RCF_MB15N405.csv'; % Selecting driver
r = 1; % Distance to from the source
eg = 2.83; % Input voltage
% Thiele/Small parameters
[Sd, fs, Vas, Qms, Bl, Re, Le, Xmax, Mms, Qts, Qes, Rms, n, dr] = TS_extract(driver, 1);
Cms = Vas / (rho * c^2 * Sd^2) % Calculating mechanical compliance
Pg=eg*Bl/(Sd*Re);
mas=rho*Sd;
% Radiation impedance uing bessel and struve functions choose
[zar,zmr]=RadImpC(Sd);
% Loop for each frequency
for k = 1:length(w)
% Transfer matrices for sub domains
tre(:, :, k) = [1, Re; 0, 1]; % voice coil resistance
tle(:, :, k) = [1, 1i*w(k)*Le; 0, 1]; % Inductance
tem(:, :, k) = [0, Bl; 1/Bl, 0]; % gyrator
tmm(:, :, k) = [1, 1i*w(k)*Mms; 0, 1]; % Mobil mass
tcm(:, :, k) = [1, 1/(1i*w(k)*Cms); 0, 1]; % Compliance
trm(:, :, k) = [1, Rms; 0, 1]; % mechanic losses
% Electromechanic chain
tdr(:, :, k) = tre(:, :, k) * tem(:, :, k) * tmm(:, :, k) * tcm(:, :, k) * trm(:, :, k);
% Calculating Iin [A] electrical current from the source from electro-mechanic
% matrix [A,B ; C,D]
t12 = tdr(1,2,k); % A
t22 = tdr(2,2,k); % B
zin = t12 / t22; % Impedance seen from source
ein = eg; % Input voltage
iin = ein / zin; % Input current
In = [ein; iin];
% Acoustic domain
tma(:, :, k) = [Sd, 0; 0, 1/Sd]; % Tranformer
tza(:, :, k) = [1, zar(k); 0, 1]; % Radiation Impedance
% Complete chain from eletrical domain to acoustical domain
Tt(:, :, k) = tre(:, :, k) * tem(:, :, k) * tmm(:, :, k) * tcm(:, :, k) * trm(:, :, k) * tma(:, :, k) * tza(:, :, k);
Ttinv = inv(Tt(:, :, k)); % Inverting matrix
Pu = Ttinv * In; % Calculating results
% Pression (pout) et vitesse acoustique (uout)
pout(k) = Pu(1); % Acoustic pressure at Sd
uout(k) = Pu(2); % Acoustic volume velocity
end
Pa = pout ./ (4 * pi * r); % Acoustic pressure at 1 [m] from the Sd
SPL = 20 * log10(abs(Pa)/ 20e-6); % Decibel SPL
G=1i.*w.*mas.*uout./Pg; % system gain
Phase_deg = unwrap(angle(G)) * 180/pi; % Phase
[fh,o,ph]=hornrespcsv("ibn4050.csv");
figure(1)
semilogx(f, SPL, 'LineWidth', 1.5, fh,o,'LineWidth',1.5);
xlabel('Frequencies (Hz)');
ylabel('SPL');
title(['SPL response']);
grid on;
figure(2)
semilogx(f, Phase_deg, 'LineWidth', 1.5,fh,ph,'LineWidth', 1.5);
xlabel('Frequencies (Hz)');
ylabel('Phase');
title(['Phase']);
grid on;
Single-stage active RIAA correction with second- or third-order Butterworth high-pass included
- By MarcelvdG
- Analogue Source
- 165 Replies
Hi all,
While thinking about ways to speed up the settling of a single-supply single-op-amp RIAA amplifier, see https://www.diyaudio.com/community/...upply-phono-preamp-design.413571/post-7702435 , I found a way to include a second- or third-order Butterworth high-pass filter. As it may be useful outside the context of single-supply circuits, I give it a separate thread.
When you just look at the topology and ignore the component values, this is a rather conventional RIAA amplifier (you could make it even more conventional by connecting R7 in parallel with C5, that doesn't matter much for the principle):
Normally, C8 is used to cause roll-off in the subsonic region and the network R7...R9, C5, C6 realizes the RIAA poles and zero. In this case, however, I use C8 to realize the lowest RIAA pole at -1/(3.18 ms) and R7 to get subsonic roll-off.
Note that C8/C5 = 1000, meaning that without the subsonic roll-off, the DC gain would be 1001, a very ordinary value for a moving-magnet amplifier (1 kHz gain roughly 40 dB).
With everything ideal, at the value of the Laplace variable s where the impedance of C8 cancels the impedance of R12, the feedback disappears and the gain goes to infinity. This means that there is a pole at exactly -1/(R12C8), so if this has to be the first RIAA pole, one needs R12C8 = 3.18 ms. It's actually 3.196 ms in the schematic, pretty close.
The disadvantage of using C8 for the first RIAA pole is that C8, which has a relatively large value, needs to be accurate to get an accurate first RIAA pole. (C8 has practically no effect on the gain at frequencies much greater than 50 Hz, so its tolerance affects deep bass, but not channel balance.) The advantage is that you can include better subsonic filtering in the loop by adding two more resistors and a capacitor.
As an intermediate step, suppose you could add an ideal inductor with a huge value between the output and the negative input of the op-amp, chosen such that it resonates with C5 at the desired subsonic roll-off frequency, and that you chose R7 such that it damps the LC circuit to a quality factor of 1/2 √2. The subsonic response would then be very close to second-order Butterworth. That's because the gain of the RIAA correction amplifier is one plus the ratio of the feedback impedance to the impedance from the negative op-amp input to ground, and that "one" is quite negligible at low frequencies. Mind you, R8 and R9 contribute to the damping of the LC circuit, but not by much. You could also choose a quality factor of 1 and design the AC coupling at the input for the same cut-off frequency. The combined response is then third-order Butterworth.
Such an ideal inductor is totally impractical, but it can be approximated with a T network consisting of two resistors with values much smaller than R7 and a capacitor to ground at the point where they are connected, see this figure:
The transfer from the voltage going into R11 to the current coming out of R10 rolls off at a first-order rate from some very low frequency onwards, like would be the case with an inductor.
I haven't found any simple exact equations for any of the values except R12C8 = 3.18 ms. In fact, I've been very lazy and just calculated approximate values for the other components, and then used a pole-zero extraction program to fine-tune the values.
Regarding those approximate calculations:
R12C8 = 3.18 ms to get the first RIAA pole at the right spot.
The DC gain would be 1 + C8/C5 without subsonic roll-off, so C8/C5 = 1000 gives you a midband gain of roughly 40 dB.
At s = -1/((R8 + R9)(C5 + C6)), the impedance of the network R8, R9, C5, C6 goes to zero and the gain of the circuit becomes 1. As a gain of 1 is pretty close to 0, this must be close to the location of the RIAA zero. That is,
(R8 + R9)(C5 + C6) ≈ 318 μs
At s = -1/((R8 + R9)C6), the impedance of the parallel connection of C6 and R8 & R9 goes to infinity. The impedance of the whole feedback network remains finite due to the other branches R7 and R11, C7, R10, but it does get pretty large. That means the second RIAA pole must be close, so we get the extra criterion
(R8 + R9)C6 ≈ 75 μs
The (theoretical) inductance L is chosen to resonate with C5 at the required subsonic roll-off frequency and R7 is chosen to get the desired quality factor. R10 and R11 get convenient values much smaller than R7 with R10 also much greater than R12. We then have
C7 = L/(R10R11)
Best regards,
Marcel
Edits:
Input RC coupling
The RCR T-network that approximates an inductor actually approximates an inductor with inductance L = R10 R11 C7 and a series resistance of R10 + R11. At low frequencies, it stops behaving inductively, it just turns into the series connection of the two resistors.
As a result, one of the zeros of the high-pass filter that are supposed to lie at s = 0 actually lies somewhere around s = -(R10 + R11)/(R10 R11 C7). For the second-order cases, I have used the first-order high-pass at the input to cover this zero by making the input RC time constant approximately equal to R10 R11 C7/(R10 + R11), or actually to a more accurate value for the displaced zero found by the LINDA pole-zero extraction program.
For the third-order case, I have used the input RC coupling to make the real pole of the third-order Butterworth response, so I couldn't use it to cover the displaced zero. I used the output RC circuit in that case, or simply did not cover the zero. The effect of the zero not being in the origin is typically only seen below 1.something Hz anyway.
There is another zero not exactly in the origin, this is related to the + 1 term in the gain expression of a non-inverting op-amp amplifier. It is so close to 0 that I decided not to bother correcting for it.
16 Hz split-supply versions
This is a version for split supply and 16 Hz cut-off frequency, see post #58, https://www.diyaudio.com/community/...rworth-high-pass-included.413649/post-7927611
For the second-order version (values in parenthesis), the input coupling capacitor C2 can be replaced with a short circuit if you don't mind when the roll-off reduces to first order below 1.3 Hz.
This is a variant with 46 dB rather than 40 dB midband gain:
Thanks to having R6 split into R6 and R0, the time constant of the input RC coupling network can be set more accurately without needing awkward values for C2. This was implicitly suggested by hbtaudio on another thread. Because of the high midband gain, the op-amp needs to have a fairly high gain-bandwidth product to get accurate RIAA correction (16 MHz gain-bandwidth product will give about -2 % error of the location of the second RIAA pole).
Finite gain-bandwidth product
See post #100, https://www.diyaudio.com/community/...rworth-high-pass-included.413649/post-7964481 , for some rough calculations on the effect of finite gain-bandwidth product of the op-amp.
From post #101 onward, Nick Sukhov points out that an amplifier with a high open-loop output impedance would result in a loop gain that depends much less on the RIAA correction circuit impedance. That's something to keep in mind when designing a discrete amplifier, you don't have the ability to choose a high open-loop output impedance when using op-amps.
Applying the subsonic filter to a discrete preamplifier based on the Hoeffelman and Meys configuration
The discussion with Nick and Chris about open-loop output impedances made me realize that the subsonic filter of this thread could be combined with a low-noise ("electrically cold") input termination resistance realized with a special feedback configuration that Dual already used in the late 1960's (CV40 phono section, see https://www.diyaudio.com/community/...o-input-load-modification.424717/post-7947176 ) and that was advocated by Hoeffelman and Meys in a 1978 AES article (Jean M. Hoeffelman and René P. Meys, "Improvements of the noise characteristics of amplifiers for magnetic transducers", Journal of the Audio Engineering Society, vol. 26, no. 12, December 1978, pages 935...939, see also Ernst H. Nordholt, "Comments on "Improvement of the noise characteristics of amplifiers for magnetic transducers"", Journal of the Audio Engineering Society, vol. 27, no. 9, September 1979, pages 680...681). The very first electrically cold resistance was made by William Spencer Percival and W. L. Horwood in 1939 as far as I know, but they used a different configuration and did not apply it to phono preamplifiers. See W. S. Percival, "An electrically "cold" resistance", The wireless engineer, vol. 16, May 1939, pages 237...240.
The schematics below show the resulting configurations. They are identical, but the left schematic is for people familiar with nullators and norators, the right schematic for people who feel more comfortable with high-gain twoports and op-amps. At frequencies well above 50 Hz, the input impedance approaches (R13 + R14)/(1 + R13/R12 + R13/R10). You can make this equal to 47 kΩ while using an R14 that is much greater than 47 kΩ, thereby reducing the thermal noise current √(4kT∆f/R) that gets injected into the input.
You can't do this with op-amps (not without floating supplies anyway) because op-amps lack the negative output that conducts a (signal) current equal but opposite to the current through the positive output. That is, you can make electrically "cold" resistances with op-amps, but not as shown here.
Document about dimensioning the circuit
The attached zip file contains a pdf document that explains step-by-step how the component values were found (section 2) and that presents a more accurate method than I have used (one that doesn't need fine-tuning with a pole-zero extraction program, section 3). It also contains a spreadsheet for the more accurate way to calculate the component values.
Deriving the expressions was a nice exercise, but I'm not at all convinced that my more accurate calculation is of any practical use. It can very easily lead to negative or complex resistances.
While thinking about ways to speed up the settling of a single-supply single-op-amp RIAA amplifier, see https://www.diyaudio.com/community/...upply-phono-preamp-design.413571/post-7702435 , I found a way to include a second- or third-order Butterworth high-pass filter. As it may be useful outside the context of single-supply circuits, I give it a separate thread.
When you just look at the topology and ignore the component values, this is a rather conventional RIAA amplifier (you could make it even more conventional by connecting R7 in parallel with C5, that doesn't matter much for the principle):
Normally, C8 is used to cause roll-off in the subsonic region and the network R7...R9, C5, C6 realizes the RIAA poles and zero. In this case, however, I use C8 to realize the lowest RIAA pole at -1/(3.18 ms) and R7 to get subsonic roll-off.
Note that C8/C5 = 1000, meaning that without the subsonic roll-off, the DC gain would be 1001, a very ordinary value for a moving-magnet amplifier (1 kHz gain roughly 40 dB).
With everything ideal, at the value of the Laplace variable s where the impedance of C8 cancels the impedance of R12, the feedback disappears and the gain goes to infinity. This means that there is a pole at exactly -1/(R12C8), so if this has to be the first RIAA pole, one needs R12C8 = 3.18 ms. It's actually 3.196 ms in the schematic, pretty close.
The disadvantage of using C8 for the first RIAA pole is that C8, which has a relatively large value, needs to be accurate to get an accurate first RIAA pole. (C8 has practically no effect on the gain at frequencies much greater than 50 Hz, so its tolerance affects deep bass, but not channel balance.) The advantage is that you can include better subsonic filtering in the loop by adding two more resistors and a capacitor.
As an intermediate step, suppose you could add an ideal inductor with a huge value between the output and the negative input of the op-amp, chosen such that it resonates with C5 at the desired subsonic roll-off frequency, and that you chose R7 such that it damps the LC circuit to a quality factor of 1/2 √2. The subsonic response would then be very close to second-order Butterworth. That's because the gain of the RIAA correction amplifier is one plus the ratio of the feedback impedance to the impedance from the negative op-amp input to ground, and that "one" is quite negligible at low frequencies. Mind you, R8 and R9 contribute to the damping of the LC circuit, but not by much. You could also choose a quality factor of 1 and design the AC coupling at the input for the same cut-off frequency. The combined response is then third-order Butterworth.
Such an ideal inductor is totally impractical, but it can be approximated with a T network consisting of two resistors with values much smaller than R7 and a capacitor to ground at the point where they are connected, see this figure:
The transfer from the voltage going into R11 to the current coming out of R10 rolls off at a first-order rate from some very low frequency onwards, like would be the case with an inductor.
I haven't found any simple exact equations for any of the values except R12C8 = 3.18 ms. In fact, I've been very lazy and just calculated approximate values for the other components, and then used a pole-zero extraction program to fine-tune the values.
Regarding those approximate calculations:
R12C8 = 3.18 ms to get the first RIAA pole at the right spot.
The DC gain would be 1 + C8/C5 without subsonic roll-off, so C8/C5 = 1000 gives you a midband gain of roughly 40 dB.
At s = -1/((R8 + R9)(C5 + C6)), the impedance of the network R8, R9, C5, C6 goes to zero and the gain of the circuit becomes 1. As a gain of 1 is pretty close to 0, this must be close to the location of the RIAA zero. That is,
(R8 + R9)(C5 + C6) ≈ 318 μs
At s = -1/((R8 + R9)C6), the impedance of the parallel connection of C6 and R8 & R9 goes to infinity. The impedance of the whole feedback network remains finite due to the other branches R7 and R11, C7, R10, but it does get pretty large. That means the second RIAA pole must be close, so we get the extra criterion
(R8 + R9)C6 ≈ 75 μs
The (theoretical) inductance L is chosen to resonate with C5 at the required subsonic roll-off frequency and R7 is chosen to get the desired quality factor. R10 and R11 get convenient values much smaller than R7 with R10 also much greater than R12. We then have
C7 = L/(R10R11)
Best regards,
Marcel
Edits:
Input RC coupling
The RCR T-network that approximates an inductor actually approximates an inductor with inductance L = R10 R11 C7 and a series resistance of R10 + R11. At low frequencies, it stops behaving inductively, it just turns into the series connection of the two resistors.
As a result, one of the zeros of the high-pass filter that are supposed to lie at s = 0 actually lies somewhere around s = -(R10 + R11)/(R10 R11 C7). For the second-order cases, I have used the first-order high-pass at the input to cover this zero by making the input RC time constant approximately equal to R10 R11 C7/(R10 + R11), or actually to a more accurate value for the displaced zero found by the LINDA pole-zero extraction program.
For the third-order case, I have used the input RC coupling to make the real pole of the third-order Butterworth response, so I couldn't use it to cover the displaced zero. I used the output RC circuit in that case, or simply did not cover the zero. The effect of the zero not being in the origin is typically only seen below 1.something Hz anyway.
There is another zero not exactly in the origin, this is related to the + 1 term in the gain expression of a non-inverting op-amp amplifier. It is so close to 0 that I decided not to bother correcting for it.
16 Hz split-supply versions
This is a version for split supply and 16 Hz cut-off frequency, see post #58, https://www.diyaudio.com/community/...rworth-high-pass-included.413649/post-7927611
For the second-order version (values in parenthesis), the input coupling capacitor C2 can be replaced with a short circuit if you don't mind when the roll-off reduces to first order below 1.3 Hz.
This is a variant with 46 dB rather than 40 dB midband gain:
Thanks to having R6 split into R6 and R0, the time constant of the input RC coupling network can be set more accurately without needing awkward values for C2. This was implicitly suggested by hbtaudio on another thread. Because of the high midband gain, the op-amp needs to have a fairly high gain-bandwidth product to get accurate RIAA correction (16 MHz gain-bandwidth product will give about -2 % error of the location of the second RIAA pole).
Finite gain-bandwidth product
See post #100, https://www.diyaudio.com/community/...rworth-high-pass-included.413649/post-7964481 , for some rough calculations on the effect of finite gain-bandwidth product of the op-amp.
From post #101 onward, Nick Sukhov points out that an amplifier with a high open-loop output impedance would result in a loop gain that depends much less on the RIAA correction circuit impedance. That's something to keep in mind when designing a discrete amplifier, you don't have the ability to choose a high open-loop output impedance when using op-amps.
Applying the subsonic filter to a discrete preamplifier based on the Hoeffelman and Meys configuration
The discussion with Nick and Chris about open-loop output impedances made me realize that the subsonic filter of this thread could be combined with a low-noise ("electrically cold") input termination resistance realized with a special feedback configuration that Dual already used in the late 1960's (CV40 phono section, see https://www.diyaudio.com/community/...o-input-load-modification.424717/post-7947176 ) and that was advocated by Hoeffelman and Meys in a 1978 AES article (Jean M. Hoeffelman and René P. Meys, "Improvements of the noise characteristics of amplifiers for magnetic transducers", Journal of the Audio Engineering Society, vol. 26, no. 12, December 1978, pages 935...939, see also Ernst H. Nordholt, "Comments on "Improvement of the noise characteristics of amplifiers for magnetic transducers"", Journal of the Audio Engineering Society, vol. 27, no. 9, September 1979, pages 680...681). The very first electrically cold resistance was made by William Spencer Percival and W. L. Horwood in 1939 as far as I know, but they used a different configuration and did not apply it to phono preamplifiers. See W. S. Percival, "An electrically "cold" resistance", The wireless engineer, vol. 16, May 1939, pages 237...240.
The schematics below show the resulting configurations. They are identical, but the left schematic is for people familiar with nullators and norators, the right schematic for people who feel more comfortable with high-gain twoports and op-amps. At frequencies well above 50 Hz, the input impedance approaches (R13 + R14)/(1 + R13/R12 + R13/R10). You can make this equal to 47 kΩ while using an R14 that is much greater than 47 kΩ, thereby reducing the thermal noise current √(4kT∆f/R) that gets injected into the input.
You can't do this with op-amps (not without floating supplies anyway) because op-amps lack the negative output that conducts a (signal) current equal but opposite to the current through the positive output. That is, you can make electrically "cold" resistances with op-amps, but not as shown here.
Document about dimensioning the circuit
The attached zip file contains a pdf document that explains step-by-step how the component values were found (section 2) and that presents a more accurate method than I have used (one that doesn't need fine-tuning with a pole-zero extraction program, section 3). It also contains a spreadsheet for the more accurate way to calculate the component values.
Deriving the expressions was a nice exercise, but I'm not at all convinced that my more accurate calculation is of any practical use. It can very easily lead to negative or complex resistances.
For Sale 2 pairs of line output transformers Lundahl LL1517
- By Groundloops
- Swap Meet
- 2 Replies
For sale 2 pairs of line output transformers Lundahl ll1517. Tested, fully working.
€75,00 per pair or €130,00 for both pairs.
€75,00 per pair or €130,00 for both pairs.
For Sale 1 pair of SB Acoustics Satori MW16P-4 mid-woofers (Europe)
- By Groundloops
- Swap Meet
- 3 Replies
1 pair of SB Acoustics Satori MW16P-4 mid-woofers. Used, working perfectly, never operated louder than 90 db/1m.
Make a fair offer ...
Make a fair offer ...
For Sale Chiurutu I2S input R2R ladder DAC PCB
Bought to experiment with and no longer required.
There's a thread about these boards here:
Requires a +5v and +12 to 15v supply. I used a silentswitcher, which was perfect for the job. I swapped the included 22uf output tants for vishay bc122 solid aluminium 33uf, just because I like these as coupling caps in other builds. There's plenty of pad options on the board to try other cap types. I only had a single 2 and 3 way block to hand, hence why they are mismatched.
£50 posted in uk, international postage at cost.
There's a thread about these boards here:
Requires a +5v and +12 to 15v supply. I used a silentswitcher, which was perfect for the job. I swapped the included 22uf output tants for vishay bc122 solid aluminium 33uf, just because I like these as coupling caps in other builds. There's plenty of pad options on the board to try other cap types. I only had a single 2 and 3 way block to hand, hence why they are mismatched.
£50 posted in uk, international postage at cost.
Ground control
- By davidbem
- Solid State
- 17 Replies
I have some ground control problems, but not Space Oddity of David Bowie. That can be a nice song in the background while reading this.
This is not my first amplifier build, so I have some knowledge and have had and solved some ground problems in the past, but this one is killing me.
First picture of amplifier layout:
Normally I build dual transformer/dual psu units. This time I decided to reduce cost and used one 600VA toroid. I have tried some different layouts of wiring, but same result.
1. With nothing connected to inputs the amplifier is absolutely silent.
2. With inputs connected together with cable - each to other, the amplifier is absolutelly silent
3. With inputs connected to source there is noise - bzzzzz. Source is Marantz amplifier with preouts - no issues with other amps.
I tried to change grounds from connector to just one point, a screw in the picture on top of capacitors, same problem.
Tried ground lift with greatz, same problem.
The solving of problem was removing the connection from common ground (the green yellow) to the eart point on top near the big resistor, where the chasis is grounded. You can see now the wire on top of the screw of common ground. If I connect this to chasis earthing point there is noise. On other amplifiers, without this wire I had noise, with this one the oposite.
What is the cause of this and can I just leave it like this. The chasis is grounded to the mains jack, but the amplifier 0V -ground is NOT.
This is not my first amplifier build, so I have some knowledge and have had and solved some ground problems in the past, but this one is killing me.
First picture of amplifier layout:
Normally I build dual transformer/dual psu units. This time I decided to reduce cost and used one 600VA toroid. I have tried some different layouts of wiring, but same result.
1. With nothing connected to inputs the amplifier is absolutely silent.
2. With inputs connected together with cable - each to other, the amplifier is absolutelly silent
3. With inputs connected to source there is noise - bzzzzz. Source is Marantz amplifier with preouts - no issues with other amps.
I tried to change grounds from connector to just one point, a screw in the picture on top of capacitors, same problem.
Tried ground lift with greatz, same problem.
The solving of problem was removing the connection from common ground (the green yellow) to the eart point on top near the big resistor, where the chasis is grounded. You can see now the wire on top of the screw of common ground. If I connect this to chasis earthing point there is noise. On other amplifiers, without this wire I had noise, with this one the oposite.
What is the cause of this and can I just leave it like this. The chasis is grounded to the mains jack, but the amplifier 0V -ground is NOT.
