Linear Audio High-voltage Delay for Tube Amplifiers V4
- By Cetla416a
- Tubes / Valves
- 7 Replies
Attachments
Help me with carbon fiber tonearm internal grounding
- By audiof001
- Analogue Source
- 42 Replies
Attachments
For Sale Hashimoto HWC-30-8 Push-Pull UL Output Transformers
- By itsikhefez
- Swap Meet
- 4 Replies
Attachments
Pass Class D
- By Nelson Pass
- Pass Labs
- 24 Replies
About 40 years ago I was inspired to make a switching amp, and the
result was a modest 20 watt/ch unit that worked quite well.
I've been asked about this several times in the last couple weeks about
the existence of a Threshold Class D amplifier, and I was inspired to
haul it out of storage and fire it up. It still measures decently if not
to the standard that Bruno has developed, but it sounded good - I
recall that Gordon Holt liked it.
Anyway, here are some pics.
result was a modest 20 watt/ch unit that worked quite well.
I've been asked about this several times in the last couple weeks about
the existence of a Threshold Class D amplifier, and I was inspired to
haul it out of storage and fire it up. It still measures decently if not
to the standard that Bruno has developed, but it sounded good - I
recall that Gordon Holt liked it.
Anyway, here are some pics.
Attachments
KEF Chorale III recap - need advice on cap values
I am recapping my KEF Chorale iii's...but the capacitor values are not any that I can find exact matches to (7uF and 5uF). What should I do here? Thank you!
Attachments
GB "long and skinny" FirstWatt J2 clone PCBs
- By JeffYoung
- Group Buys
- 538 Replies
I'm producing a set of J2 boards for myself and thought I'd do a group buy to increase volume and reduce costs a bit.
Boards are the "long and skinny" format for FirstWatt-style enclosures with a flanged heat-sink. (Boards are 14" long with 2" hole spacing.) LTP layout is for dual SJ109s (although quad SJ74s will go in the same holes).
Boardhouse will be EuroCircuits. 35uM copper (~60uM after plating). Cost will be between €15 and €25 per channel (depending on volume) + shipping. Cost includes EU VAT.
Schematic, BOM, etc. in post #15 (GB "long and skinny" FirstWatt J2 clone PCBs).
Boards are the "long and skinny" format for FirstWatt-style enclosures with a flanged heat-sink. (Boards are 14" long with 2" hole spacing.) LTP layout is for dual SJ109s (although quad SJ74s will go in the same holes).
Boardhouse will be EuroCircuits. 35uM copper (~60uM after plating). Cost will be between €15 and €25 per channel (depending on volume) + shipping. Cost includes EU VAT.
Schematic, BOM, etc. in post #15 (GB "long and skinny" FirstWatt J2 clone PCBs).
Attachments
My OP-AMP rolling adventures
Amplifier build is here:
https://www.diyaudio.com/community/threads/finished-my-class-d-build-sounds-amazing.401362/
-----------------------
Very happy with the sound of the amp, but I wanted to see if a positive impact could be made with OP-AMP changes.
Supplied with the zero zone TPA3255 boards I used are the JRC 5532DD OP-AMPS. Despite being nothing special they are a great match for the amplifier boards. Detail and separation are fantastic, I was quite happy with them.
After posting here and doing a bit of reading I decided to try the Texas Instruments OPA1656. Quite well regarded and made by TI who also make the 3255 chip contained in the amp boards. I was expecting good things.
First kick in the teeth was the eBay seller, or rather the packaging. Just a thin plastic sandwich bag. Inevitably one of the pair arrived with half its legs smashed off! The seller refused to replace, only refund. So I bought a replacement from another seller. Same awful packaging, this time bent legs which I carefully straightened.
They lasted 20 minutes in my system before the JRC's went back in. They sounded like pressing a loudness button with low and high end frequencies being boosted leaving the midrange (and thus a lot of detail) muffled and drowning. Soundstage was ruined, with the system now sounding like I had draped a duvet Infront of the speaker's .
Next up Burson Audio 'Vivid'
The first nice thing is the packaging. Nice and safe, no bent of missing legs. Off to a good start over the 1656!
These are quite large both in footprint and height. Not a problem on my boards but it could be for some.
Initial thoughts, bass was the first thing I noticed. I run my system with a pair of wharfedale Diamond 9.1's, which are fantastic speakers in my opinion, but somewhat lacking in the bass department. That said I usually listen at low volumes and I have spent many hours enjoying my system how it was.
I wasn't sure on the increase in bass that the Burson Vivids created at first. But continued testing. All of the detail and separation this system did so well was still present with an unexpected added bonus once in a blue moon.... the odd twang of a guitar at the end of a riff during Fleetwood Macs Rumours (SACD) would really pop and travel. Almost like in a surround system. The FOO FIGHTERS greatest hits provided no real changes over the JRC's apart from the previously mentioned increase in bass which after briefly swapping back to the JRC's and playing the same song again, I must say was now proving to be an improvement.
Now a volume increase to medium levels and some Bonobo - Fragments (24bit). A very atmospheric album with lots of little details like triangle twangs and sparkles. The Bursons did well here. Thankfully they left the soundstage alone just adding a touch of bass and the odd sparkle. I found myself waiting for these sparkly almost surround sound like moments. Nice.
After a couple of hours the increased bass was now a feature that I enjoy and having swapped back and to the JRC's a few times now, I need in my life.
For anyone who hasn't swapped around OP-AMPS before, we are not talking huge changes here. If I had to put a figure on it, somewhere between 5 & 10% (certainly with the opamps I used) subtle but notible.
In short I rate the Bursons, they will be staying in.
https://www.diyaudio.com/community/threads/finished-my-class-d-build-sounds-amazing.401362/
-----------------------
Very happy with the sound of the amp, but I wanted to see if a positive impact could be made with OP-AMP changes.
Supplied with the zero zone TPA3255 boards I used are the JRC 5532DD OP-AMPS. Despite being nothing special they are a great match for the amplifier boards. Detail and separation are fantastic, I was quite happy with them.
After posting here and doing a bit of reading I decided to try the Texas Instruments OPA1656. Quite well regarded and made by TI who also make the 3255 chip contained in the amp boards. I was expecting good things.
First kick in the teeth was the eBay seller, or rather the packaging. Just a thin plastic sandwich bag. Inevitably one of the pair arrived with half its legs smashed off! The seller refused to replace, only refund. So I bought a replacement from another seller. Same awful packaging, this time bent legs which I carefully straightened.
They lasted 20 minutes in my system before the JRC's went back in. They sounded like pressing a loudness button with low and high end frequencies being boosted leaving the midrange (and thus a lot of detail) muffled and drowning. Soundstage was ruined, with the system now sounding like I had draped a duvet Infront of the speaker's .
Next up Burson Audio 'Vivid'
The first nice thing is the packaging. Nice and safe, no bent of missing legs. Off to a good start over the 1656!
These are quite large both in footprint and height. Not a problem on my boards but it could be for some.
Initial thoughts, bass was the first thing I noticed. I run my system with a pair of wharfedale Diamond 9.1's, which are fantastic speakers in my opinion, but somewhat lacking in the bass department. That said I usually listen at low volumes and I have spent many hours enjoying my system how it was.
I wasn't sure on the increase in bass that the Burson Vivids created at first. But continued testing. All of the detail and separation this system did so well was still present with an unexpected added bonus once in a blue moon.... the odd twang of a guitar at the end of a riff during Fleetwood Macs Rumours (SACD) would really pop and travel. Almost like in a surround system. The FOO FIGHTERS greatest hits provided no real changes over the JRC's apart from the previously mentioned increase in bass which after briefly swapping back to the JRC's and playing the same song again, I must say was now proving to be an improvement.
Now a volume increase to medium levels and some Bonobo - Fragments (24bit). A very atmospheric album with lots of little details like triangle twangs and sparkles. The Bursons did well here. Thankfully they left the soundstage alone just adding a touch of bass and the odd sparkle. I found myself waiting for these sparkly almost surround sound like moments. Nice.
After a couple of hours the increased bass was now a feature that I enjoy and having swapped back and to the JRC's a few times now, I need in my life.
For anyone who hasn't swapped around OP-AMPS before, we are not talking huge changes here. If I had to put a figure on it, somewhere between 5 & 10% (certainly with the opamps I used) subtle but notible.
In short I rate the Bursons, they will be staying in.
Hello to all , I just joined up
- By leonardo789
- Introductions
- 7 Replies
Nice to join up and hear from fellows who are enjoying the hi-fi journey .I have made a few speakers and currently liking and and admiring the SB acoustics 20frpc30 fullranger .
How critical is achieving perfect RIAA equalization?
- By SomeJoe
- Electronic Design
- 25 Replies
Hoping I'm not opening a Pandora's box but here goes 😛
I'm still in the process of designing my Rega Elicit phono clone and been wondering about this. In one of my books by a reputable author, he describes how to achieve perfect network component values for the resistors and capacitors by paralleling each component with a triming one, with the requirement that the ratio of the main and trim values be greater than the ratio of their tolerances. Of course the value measurements have to be made with a precision bridge. This is particularly helpful if one has come up with "awkward" values outside of the standard series, which is not my case.
I don't have a bridge and honestly I think it might be overkill to do this. Reason is, one of the components of the network is the output impedance of the gain stage which I believe will have worse tolerance than the 0.1% resistors and 1% capacitors the author suggests unless output transistors and their associated circuit components are also selected to be within these tolerances. And I seriously doubt manufacturers go to these lengths.
But out of curiosity I simulated my design with 1% variation factors for the passive components of both passive and active eq filters; see frequency response output plot below for 27 seven different combinations. Yes the frequency response is less flat but the maximum deviation from the theorical values is on the order of 0.3dB peak-peak for the extremes. Can this difference really be heard?
Just wondering if I need to go to the trouble of adding space for extra trim components on my PCB. I might do it for resistors as this is not that complicated but for the polystyrene caps I'm not sure as there are large variations in footprints of the parts I can get my hands on off ebay.
Thanks in advance for any insights.
-Joris
I'm still in the process of designing my Rega Elicit phono clone and been wondering about this. In one of my books by a reputable author, he describes how to achieve perfect network component values for the resistors and capacitors by paralleling each component with a triming one, with the requirement that the ratio of the main and trim values be greater than the ratio of their tolerances. Of course the value measurements have to be made with a precision bridge. This is particularly helpful if one has come up with "awkward" values outside of the standard series, which is not my case.
I don't have a bridge and honestly I think it might be overkill to do this. Reason is, one of the components of the network is the output impedance of the gain stage which I believe will have worse tolerance than the 0.1% resistors and 1% capacitors the author suggests unless output transistors and their associated circuit components are also selected to be within these tolerances. And I seriously doubt manufacturers go to these lengths.
But out of curiosity I simulated my design with 1% variation factors for the passive components of both passive and active eq filters; see frequency response output plot below for 27 seven different combinations. Yes the frequency response is less flat but the maximum deviation from the theorical values is on the order of 0.3dB peak-peak for the extremes. Can this difference really be heard?
Just wondering if I need to go to the trouble of adding space for extra trim components on my PCB. I might do it for resistors as this is not that complicated but for the polystyrene caps I'm not sure as there are large variations in footprints of the parts I can get my hands on off ebay.
Thanks in advance for any insights.
-Joris
The Journey of DIY No-Feedback Class D Amplifier (3). Subtitle: Simulation Results and Reflections on the Circuit Design
After finalizing the overall design, simulations are used to select specific components and determine resistor and capacitor values, followed by circuit board manufacturing for validation and iteration to achieve the initial design goals.
The Σ-Δ (Sigma-Delta) modulation method was chosen for the following reasons:
1，A fixed-frequency PWM modulator has almost zero Power Supply Rejection Ratio(PSRR) in its output stage, meaning it has no ability to suppress power noise. This creates significant challenges for power supply design, as any noise from the power supply directly converts into audio noise and reaches the headphones. In contrast, the Σ-Δ modulator offers some level of power supply noise rejection, greatly reducing the burden on power supply design.
2，Fixed-frequency PWM modulators use a standard triangular wave (blue waveform in Figure 1(a)), which contains a lot of high-frequency components. However, the integrator-generated waveforms from multi-stage integrators (green waveform in Figure 1(a)) resemble a triangular shape but with fewer high-frequency components and faster high-frequency roll-off (as shown in Figure 1(b), with blue representing the standard triangular wave's spectrum and green representing the multi-stage integrator waveform's spectrum). In an amplifier with the same bandwidth, the multi-stage integrator waveform results in lower distortion, and the PDM modulation signal's edge jitter (rise and fall times) is smaller, leading to overall lower noise and distortion of audio signal.
(a) (b)
Figure 1, Triangle Wave, Higher-Order Integral Wave, and Their Spectrum
3，The Σ-Δ modulator can use a differential integrator circuit, which outputs a pair of equal-amplitude, opposite-polarity integrator waveforms (see Figure 2(b)) to the comparator. In contrast, the fixed-frequency PWM modulator sends a triangular wave and input signal to the comparator (Figure 2(a) shows the comparator input waveform of the fixed-frequency PWM modulator). It's clear that the slope (rate of change) at the intersection points of these two sets of input waveforms differs significantly. The differential integrator waveforms have a much steeper slope at the intersection points compared to the PWM modulator’s waveforms. This means that for the same noise level, the edge jitter of the comparator output is smaller, thus improving the overall amplifier performance. Attentive readers might have figured out that the slope at the signal crossover points of the higher-order integrator waveform should be much steeper than that of the standard triangular wave, which could be one of the reasons why the higher-order integrator performs better than the standard triangular wave.
(a) (b)
Figure 2， Comparator Input Signals of PWM and PDM Modulators
4，Compared to fixed-frequency PWM modulators, Σ-Δ modulators offer greater flexibility and room for improvement in design. For example, a Schmitt trigger mechanism can be added to the comparator in a Σ-Δ modulator to improve the stability of the PDM pulse waveform edges.
The discussion so far has focused on reducing edge jitter in the PDM pulse signal. In Class D amplifiers, the pulse signal is central to the entire system; only precise pulses can result in low noise and low THD distortion. The edges of these pulses are crucial. If edge jitter is unrelated to the input signal, it generates noise, which, while having a smaller effect on distortion, can increase the amplifier’s background noise. If edge jitter is related to the input signal—such as when the triangular wave lacks linearity—the jitter (early or late edges) will correlate with the input signal amplitude, causing nonlinear distortion and resulting in higher THD.
Through the improvements discussed above, Σ-Δ modulation can potentially achieve lower distortion. The next step is to compare performance through simulation to select different components and parameters. This is a complex and tedious process, as components interact with each other, and changes to one typically require adjustments to related parameters. I'll use a key example to explain this further.
The precision of the modulated pulse signal comes from the accuracy of the multi-order integrator signals. The amplitude and frequency of these integrator waveforms are affected by many factors, such as the zero and pole distribution in the integrator network, the depth of the Schmitt trigger in the comparator, and the oscillation control resistor. For simplicity, Figure 3 shows a basic schematic of a multi-order half-bridge Σ-Δ modulator with a Schmitt trigger. The ratio of the Schmitt trigger resistor influences the modulation frequency, and the equivalent capacitance of the high-order integrator also affects the modulation frequency. These factors further shape the integrator waveform and its amplitude.
The accuracy of the integrator waveform has already been analyzed, but its amplitude is equally important, requiring a balance between the output capability of the integrator and the performance of the comparator. In general, a larger signal fed into the comparator results in more precise outputs, but an overly large output from the integrator can cause signal distortion.
Figure 3， Multistage Half-Bridge Sigma-Delta Modulator with Schmitt Trigger Mechanism
The second important aspect is the selection of the output stage, where the industry has reached a consensus: when cost requirements are not stringent, a full-bridge configuration is typically used. The advantages of the full-bridge mode include the ability to partially cancel even-order harmonics and suppress the back current of the output filter inductor, effectively reducing the power supply fluctuations in the output stage. This is particularly beneficial for Class D amplifiers with poor power supply rejection ratio (PSRR).
The output filter is also critical, especially for Class D amplifiers using PDM modulation. Since this design uses a modulation frequency above 2 MHz, there is more flexibility in choosing the low-pass filter’s cutoff frequency. This brings significant benefits: first, a higher cutoff frequency can be chosen, so the frequency response changes with different loads won't significantly impact the amplifier’s effective bandwidth. Second, since smaller inductors are used, they provide better linearity at the same size, reducing the distortion impact on the amplifier. During simulation, it was also observed that a multi-order output filter could reduce in-band noise, although there is no theoretical explanation for this yet. I welcome any interested parties to discuss this further.
The power supply solution is equally important. This design uses a single 3000mAh lithium battery, so a step-up boost converter is required. A major issue is that the amplifier operates at over 2 MHz, which is much higher than the frequency of commonly available boost converters. During a single PWM cycle, the supply voltage may drop due to the amplifier’s high current draw, which even the best boost chips struggle to handle in such cases.
After multiple experiments, both the pre-amp and output stages ended up using a two-stage single power supply solution. The pre-amp power (for the integrator and comparator) uses a combination of a boost circuit and a Class A power supply. The Class A supply uses a high-precision reference regulator with voltage accuracy of 0.1% and power supply noise as low as a few microvolts. Additionally, the midpoint reference voltage uses the same precision reference chip, not only improving voltage accuracy but also ensuring the integrator circuit operates symmetrically, optimizing open-loop performance.
Originally, I planned to use the same design for the output stage power supply, but due to the need to drive headphones with varying impedance and sensitivity, the Class A power supply’s efficiency was too low, which affected battery life. In the end, I chose a step-up circuit combined with a high-precision, high-current LDO and large capacitors for the output stage.
Interestingly, switching from a Class A supply to an LDO + large capacitors for the output stage resulted in virtually no subjective change in sound quality. However, changing the pre-amp from an LDO to a Class A power supply led to a significant improvement in sound. The high frequencies became more transparent, the low frequencies were stronger and more extended, and the soundstage was more stable. This change was quite noticeable, without the need for careful listening. Could this be due to the higher precision in both supply and bias voltages? I would love to hear thoughts from those interested in discussing this. The first version of the pre-amp, which was discarded, used a voltage divider for the bias reference, as shown in Figure 4, with resistor accuracy of 0.1%.
Figure 4， Circuit for Generating the Bias Voltage in the Integrator
There are many other crucial details in the amplifier design and debugging process that cannot be overlooked, and I will discuss them further in appropriate contexts.
Figure 5 shows the Spice simulation results for the background noise. Within an effective bandwidth of 35 kHz, the highest background noise is as low as -170 dBV, with the low-frequency section reaching nearly -180 dBV.
Figure 5，Zero Input Circuit Configuration (Background Noise) Simulation Results
Figure 6 shows the simulation results with a 1kHz input signal. No harmonic distortion related to the 1kHz frequency was observed. However, compared to the zero-input results in Figure 5, the background noise increased by about 10dB, reaching approximately -170dBV. This frequency-independent distortion is a typical occurrence in such circuits.
Figure 6， Single Sine Wave Input Circuit Configuration Simulation Results
Figure 7 presents the amplifier simulation results based on the previous discussions. The modulation frequency is around 2.3 MHz, with an input signal peak-to-peak value of about 196 mV and an output peak-to-peak value of around 4000 mV, resulting in an overall gain of approximately 26 dB. The input signal consists of three frequencies: 500 Hz, 5 kHz, and 20 kHz. The amplifier's background noise level is about -167 dBV, which is 3 dB higher than the -170 dBV observed in the 1 kHz sine wave test shown in Figure 6, likely due to frequency-independent distortion causing this noise floor increase.
During the simulation, it was observed that different multi-stage integrator network configurations affect the spectral characteristics of frequency-independent distortion. By adjusting the integrator network, it is possible to reduce low-frequency distortion at the cost of increasing high-frequency distortion. In this design, I chose to balance the frequency-independent distortion across the spectrum, leading to a flat noise floor.
In reality, there are other approaches. One could adjust the integrator network to minimize the total or weighted power of frequency-independent noise within the effective bandwidth, without prioritizing flatness. My choice for a flat noise floor was based on the fact that amplifiers typically exhibit more distortion in high frequencies compared to low frequencies, whether it is frequency-dependent or independent. A DIY amplifier with a flat noise floor might offer a different subjective listening experience.
Given the multi-frequency input, the amplifier's total distortion is primarily composed of harmonic and intermodulation distortion. The simulation results show that the highest harmonic distortion reaches about -156 dBV, with the minimum signal-to-harmonic ratio being 152 dB. These results demonstrate excellent noise suppression and distortion control, achieving the expected high-performance standards, at least based on the simulation outcomes.
Figure 7，Multi-Sine Wave Input Circuit Configuration Simulation Results
The following lists the different distortion frequencies and their amplitudes generated from the simulation, for reference by those interested:
For those who might not be familiar with these simulation figures and find them hard to interpret, it's worth noting that if you were to input a true 24-bit audio source into the amplifier, the output would be virtually undistorted. The harmonic distortion is more than 7 dB lower than the 24-bit quantization noise, making it effectively distortion-free.
In addition, I added negative feedback to the circuit from Figure 7 and adjusted the overall gain to 20 dB, 6 dB lower than without feedback. Observing the results in Figure 8, no frequency-dependent harmonics were observed, but the overall noise floor increased by about 1 dB compared to the no-feedback case.
This phenomenon may differ from traditional Class B amplifiers. Class D amplifiers, due to their modulation frequency, generate out-of-band noise. When negative feedback is introduced, this out-of-band noise can also be fed back to the input. While negative feedback effectively suppresses harmonic distortion caused by nonlinearity, the feedback of out-of-band noise may increase frequency-independent distortion, thus raising the overall noise floor.
Although this analysis partially explains the rise in noise floor, it has not been rigorously validated and is provided for reference only. If I gather further experimental data or validation, I will share it separately in the future.
Figure 8， Simulation Results of Multi-Sine Wave Input Circuit Scheme with Global Feedback
Additionally, when observing the operating frequency of the amplifier, it becomes apparent that as the peak value of the input signal increases, the divergence of the main frequency enhances, while the peak itself becomes lower. This phenomenon is a characteristic of the sigma-delta self-oscillation architecture, providing significant advantages in electromagnetic compatibility (EMC). Since the self-oscillation frequency is not fixed, it dynamically changes under different load and signal conditions, dispersing EMI (electromagnetic interference) energy rather than concentrating it at a specific frequency. This reduces interference with other devices, making sigma-delta amplifiers relatively superior in EMC performance in practical applications.
The following text is excerpted from "Audio Power AMP Design Handbook 4th edition" by Douglas Self, which discusses the increasing distortion characteristics of traditional Class B amplifiers at higher frequencies. Although this book was published in 2006, and advancements in technology may have significantly improved the high-frequency distortion characteristics of current amplifiers, this fundamental principle still holds unless there are breakthrough developments.
From the simulation results, Class D amplifiers also adhere to this pattern, but their high-frequency distortion is only a few dB higher than the low-frequency distortion. Given the overall low distortion rates, this difference can generally be ignored.
Another noteworthy aspect is that the amplifier's phase response tends to be linear. As we know, the human ear is not very sensitive to phase response in sound. Therefore, it raises an interesting question: do linear-phase amplifiers produce different subjective listening experiences? If anyone has personal experiences or insights on this, it would be great to share and discuss!
Based on a simple calculation of the simulated distortion effects, the amplifier's THD value can reach approximately 0.0000025% with the minimal SNR is 152dB. Considering that the harmonic components are several dB above the noise floor, the THD+N value should be in the same range. This simulation result seems almost unrealistically good, so I conducted multiple verification simulations and cross-checked with actual test results, ultimately confirming that the simulation results have a certain degree of credibility. The verification approach was as follows:
Of course, the performance results from the simulation represent only a theoretical upper limit; the actual amplifier's performance may degrade to varying degrees due to various factors. A good simulation result is merely a starting point, and the process of building the amplifier is essentially a struggle against these degradation factors.
Writing the threads has been particularly challenging for me. I’ve tried to clearly describe the simulation conditions at the time, but the reality is far more complex than what I've expressed here. If there are any unclear or incorrect points, I welcome further discussion. For now, I will share the results of my two major version of DIY AMP and some subjective listening experiences, so stay tuned!
The Σ-Δ (Sigma-Delta) modulation method was chosen for the following reasons:
1，A fixed-frequency PWM modulator has almost zero Power Supply Rejection Ratio(PSRR) in its output stage, meaning it has no ability to suppress power noise. This creates significant challenges for power supply design, as any noise from the power supply directly converts into audio noise and reaches the headphones. In contrast, the Σ-Δ modulator offers some level of power supply noise rejection, greatly reducing the burden on power supply design.
2，Fixed-frequency PWM modulators use a standard triangular wave (blue waveform in Figure 1(a)), which contains a lot of high-frequency components. However, the integrator-generated waveforms from multi-stage integrators (green waveform in Figure 1(a)) resemble a triangular shape but with fewer high-frequency components and faster high-frequency roll-off (as shown in Figure 1(b), with blue representing the standard triangular wave's spectrum and green representing the multi-stage integrator waveform's spectrum). In an amplifier with the same bandwidth, the multi-stage integrator waveform results in lower distortion, and the PDM modulation signal's edge jitter (rise and fall times) is smaller, leading to overall lower noise and distortion of audio signal.
(a) (b)
Figure 1, Triangle Wave, Higher-Order Integral Wave, and Their Spectrum
3，The Σ-Δ modulator can use a differential integrator circuit, which outputs a pair of equal-amplitude, opposite-polarity integrator waveforms (see Figure 2(b)) to the comparator. In contrast, the fixed-frequency PWM modulator sends a triangular wave and input signal to the comparator (Figure 2(a) shows the comparator input waveform of the fixed-frequency PWM modulator). It's clear that the slope (rate of change) at the intersection points of these two sets of input waveforms differs significantly. The differential integrator waveforms have a much steeper slope at the intersection points compared to the PWM modulator’s waveforms. This means that for the same noise level, the edge jitter of the comparator output is smaller, thus improving the overall amplifier performance. Attentive readers might have figured out that the slope at the signal crossover points of the higher-order integrator waveform should be much steeper than that of the standard triangular wave, which could be one of the reasons why the higher-order integrator performs better than the standard triangular wave.
(a) (b)
Figure 2， Comparator Input Signals of PWM and PDM Modulators
4，Compared to fixed-frequency PWM modulators, Σ-Δ modulators offer greater flexibility and room for improvement in design. For example, a Schmitt trigger mechanism can be added to the comparator in a Σ-Δ modulator to improve the stability of the PDM pulse waveform edges.
The discussion so far has focused on reducing edge jitter in the PDM pulse signal. In Class D amplifiers, the pulse signal is central to the entire system; only precise pulses can result in low noise and low THD distortion. The edges of these pulses are crucial. If edge jitter is unrelated to the input signal, it generates noise, which, while having a smaller effect on distortion, can increase the amplifier’s background noise. If edge jitter is related to the input signal—such as when the triangular wave lacks linearity—the jitter (early or late edges) will correlate with the input signal amplitude, causing nonlinear distortion and resulting in higher THD.
Through the improvements discussed above, Σ-Δ modulation can potentially achieve lower distortion. The next step is to compare performance through simulation to select different components and parameters. This is a complex and tedious process, as components interact with each other, and changes to one typically require adjustments to related parameters. I'll use a key example to explain this further.
The precision of the modulated pulse signal comes from the accuracy of the multi-order integrator signals. The amplitude and frequency of these integrator waveforms are affected by many factors, such as the zero and pole distribution in the integrator network, the depth of the Schmitt trigger in the comparator, and the oscillation control resistor. For simplicity, Figure 3 shows a basic schematic of a multi-order half-bridge Σ-Δ modulator with a Schmitt trigger. The ratio of the Schmitt trigger resistor influences the modulation frequency, and the equivalent capacitance of the high-order integrator also affects the modulation frequency. These factors further shape the integrator waveform and its amplitude.
The accuracy of the integrator waveform has already been analyzed, but its amplitude is equally important, requiring a balance between the output capability of the integrator and the performance of the comparator. In general, a larger signal fed into the comparator results in more precise outputs, but an overly large output from the integrator can cause signal distortion.
Figure 3， Multistage Half-Bridge Sigma-Delta Modulator with Schmitt Trigger Mechanism
The second important aspect is the selection of the output stage, where the industry has reached a consensus: when cost requirements are not stringent, a full-bridge configuration is typically used. The advantages of the full-bridge mode include the ability to partially cancel even-order harmonics and suppress the back current of the output filter inductor, effectively reducing the power supply fluctuations in the output stage. This is particularly beneficial for Class D amplifiers with poor power supply rejection ratio (PSRR).
The output filter is also critical, especially for Class D amplifiers using PDM modulation. Since this design uses a modulation frequency above 2 MHz, there is more flexibility in choosing the low-pass filter’s cutoff frequency. This brings significant benefits: first, a higher cutoff frequency can be chosen, so the frequency response changes with different loads won't significantly impact the amplifier’s effective bandwidth. Second, since smaller inductors are used, they provide better linearity at the same size, reducing the distortion impact on the amplifier. During simulation, it was also observed that a multi-order output filter could reduce in-band noise, although there is no theoretical explanation for this yet. I welcome any interested parties to discuss this further.
The power supply solution is equally important. This design uses a single 3000mAh lithium battery, so a step-up boost converter is required. A major issue is that the amplifier operates at over 2 MHz, which is much higher than the frequency of commonly available boost converters. During a single PWM cycle, the supply voltage may drop due to the amplifier’s high current draw, which even the best boost chips struggle to handle in such cases.
After multiple experiments, both the pre-amp and output stages ended up using a two-stage single power supply solution. The pre-amp power (for the integrator and comparator) uses a combination of a boost circuit and a Class A power supply. The Class A supply uses a high-precision reference regulator with voltage accuracy of 0.1% and power supply noise as low as a few microvolts. Additionally, the midpoint reference voltage uses the same precision reference chip, not only improving voltage accuracy but also ensuring the integrator circuit operates symmetrically, optimizing open-loop performance.
Originally, I planned to use the same design for the output stage power supply, but due to the need to drive headphones with varying impedance and sensitivity, the Class A power supply’s efficiency was too low, which affected battery life. In the end, I chose a step-up circuit combined with a high-precision, high-current LDO and large capacitors for the output stage.
Interestingly, switching from a Class A supply to an LDO + large capacitors for the output stage resulted in virtually no subjective change in sound quality. However, changing the pre-amp from an LDO to a Class A power supply led to a significant improvement in sound. The high frequencies became more transparent, the low frequencies were stronger and more extended, and the soundstage was more stable. This change was quite noticeable, without the need for careful listening. Could this be due to the higher precision in both supply and bias voltages? I would love to hear thoughts from those interested in discussing this. The first version of the pre-amp, which was discarded, used a voltage divider for the bias reference, as shown in Figure 4, with resistor accuracy of 0.1%.
Figure 4， Circuit for Generating the Bias Voltage in the Integrator
There are many other crucial details in the amplifier design and debugging process that cannot be overlooked, and I will discuss them further in appropriate contexts.
Figure 5 shows the Spice simulation results for the background noise. Within an effective bandwidth of 35 kHz, the highest background noise is as low as -170 dBV, with the low-frequency section reaching nearly -180 dBV.
Figure 5，Zero Input Circuit Configuration (Background Noise) Simulation Results
Figure 6 shows the simulation results with a 1kHz input signal. No harmonic distortion related to the 1kHz frequency was observed. However, compared to the zero-input results in Figure 5, the background noise increased by about 10dB, reaching approximately -170dBV. This frequency-independent distortion is a typical occurrence in such circuits.
Figure 6， Single Sine Wave Input Circuit Configuration Simulation Results
Figure 7 presents the amplifier simulation results based on the previous discussions. The modulation frequency is around 2.3 MHz, with an input signal peak-to-peak value of about 196 mV and an output peak-to-peak value of around 4000 mV, resulting in an overall gain of approximately 26 dB. The input signal consists of three frequencies: 500 Hz, 5 kHz, and 20 kHz. The amplifier's background noise level is about -167 dBV, which is 3 dB higher than the -170 dBV observed in the 1 kHz sine wave test shown in Figure 6, likely due to frequency-independent distortion causing this noise floor increase.
During the simulation, it was observed that different multi-stage integrator network configurations affect the spectral characteristics of frequency-independent distortion. By adjusting the integrator network, it is possible to reduce low-frequency distortion at the cost of increasing high-frequency distortion. In this design, I chose to balance the frequency-independent distortion across the spectrum, leading to a flat noise floor.
In reality, there are other approaches. One could adjust the integrator network to minimize the total or weighted power of frequency-independent noise within the effective bandwidth, without prioritizing flatness. My choice for a flat noise floor was based on the fact that amplifiers typically exhibit more distortion in high frequencies compared to low frequencies, whether it is frequency-dependent or independent. A DIY amplifier with a flat noise floor might offer a different subjective listening experience.
Given the multi-frequency input, the amplifier's total distortion is primarily composed of harmonic and intermodulation distortion. The simulation results show that the highest harmonic distortion reaches about -156 dBV, with the minimum signal-to-harmonic ratio being 152 dB. These results demonstrate excellent noise suppression and distortion control, achieving the expected high-performance standards, at least based on the simulation outcomes.
Figure 7，Multi-Sine Wave Input Circuit Configuration Simulation Results
The following lists the different distortion frequencies and their amplitudes generated from the simulation, for reference by those interested:
- 1.5 kHz, -164.3 dBV
- 4 kHz, -159.5 dBV
- 9.5 kHz, -159.1 dBV
- 10 kHz, -160.7 dBV
- 10.5 kHz, -160.2 dBV
- 14.5 kHz, -157 dBV
- 15.5 kHz, -157.0 dBV
- 19 kHz, -157.1 dBV
For those who might not be familiar with these simulation figures and find them hard to interpret, it's worth noting that if you were to input a true 24-bit audio source into the amplifier, the output would be virtually undistorted. The harmonic distortion is more than 7 dB lower than the 24-bit quantization noise, making it effectively distortion-free.
In addition, I added negative feedback to the circuit from Figure 7 and adjusted the overall gain to 20 dB, 6 dB lower than without feedback. Observing the results in Figure 8, no frequency-dependent harmonics were observed, but the overall noise floor increased by about 1 dB compared to the no-feedback case.
This phenomenon may differ from traditional Class B amplifiers. Class D amplifiers, due to their modulation frequency, generate out-of-band noise. When negative feedback is introduced, this out-of-band noise can also be fed back to the input. While negative feedback effectively suppresses harmonic distortion caused by nonlinearity, the feedback of out-of-band noise may increase frequency-independent distortion, thus raising the overall noise floor.
Although this analysis partially explains the rise in noise floor, it has not been rigorously validated and is provided for reference only. If I gather further experimental data or validation, I will share it separately in the future.
Figure 8， Simulation Results of Multi-Sine Wave Input Circuit Scheme with Global Feedback
Additionally, when observing the operating frequency of the amplifier, it becomes apparent that as the peak value of the input signal increases, the divergence of the main frequency enhances, while the peak itself becomes lower. This phenomenon is a characteristic of the sigma-delta self-oscillation architecture, providing significant advantages in electromagnetic compatibility (EMC). Since the self-oscillation frequency is not fixed, it dynamically changes under different load and signal conditions, dispersing EMI (electromagnetic interference) energy rather than concentrating it at a specific frequency. This reduces interference with other devices, making sigma-delta amplifiers relatively superior in EMC performance in practical applications.
The following text is excerpted from "Audio Power AMP Design Handbook 4th edition" by Douglas Self, which discusses the increasing distortion characteristics of traditional Class B amplifiers at higher frequencies. Although this book was published in 2006, and advancements in technology may have significantly improved the high-frequency distortion characteristics of current amplifiers, this fundamental principle still holds unless there are breakthrough developments.
From the simulation results, Class D amplifiers also adhere to this pattern, but their high-frequency distortion is only a few dB higher than the low-frequency distortion. Given the overall low distortion rates, this difference can generally be ignored.
Another noteworthy aspect is that the amplifier's phase response tends to be linear. As we know, the human ear is not very sensitive to phase response in sound. Therefore, it raises an interesting question: do linear-phase amplifiers produce different subjective listening experiences? If anyone has personal experiences or insights on this, it would be great to share and discuss!
Based on a simple calculation of the simulated distortion effects, the amplifier's THD value can reach approximately 0.0000025% with the minimal SNR is 152dB. Considering that the harmonic components are several dB above the noise floor, the THD+N value should be in the same range. This simulation result seems almost unrealistically good, so I conducted multiple verification simulations and cross-checked with actual test results, ultimately confirming that the simulation results have a certain degree of credibility. The verification approach was as follows:
- Device SPICE Model Verification: Each device's SPICE model was individually verified by building its recommended test circuit using the model and comparing the results with those provided by the manufacturer. The simulation results matched the manufacturer's data closely, confirming the accuracy of the models used.
- Comparison of Simulation and Measured Parameters: Key parameters of the fabricated circuit (such as the amplitude and shape of the integral waveform, the amplifier's oscillation frequency, and gain) were measured and compared with simulation results, with discrepancies generally kept within 10%. This directly demonstrates the reliability of the simulated circuit.
Of course, the performance results from the simulation represent only a theoretical upper limit; the actual amplifier's performance may degrade to varying degrees due to various factors. A good simulation result is merely a starting point, and the process of building the amplifier is essentially a struggle against these degradation factors.
Writing the threads has been particularly challenging for me. I’ve tried to clearly describe the simulation conditions at the time, but the reality is far more complex than what I've expressed here. If there are any unclear or incorrect points, I welcome further discussion. For now, I will share the results of my two major version of DIY AMP and some subjective listening experiences, so stay tuned!
For Sale Pair Tektronix P6111B 200mHz Probes
SOLD, thank you.
Excellent condition pair of Tektronix P6111B 200mHz 10X probes, complete.
Shipping is included (only to CONUS).
$46 for both, shipped, including PP G&S fee, or $44 if paid by F&F.
Thank you.
Excellent condition pair of Tektronix P6111B 200mHz 10X probes, complete.
Shipping is included (only to CONUS).
$46 for both, shipped, including PP G&S fee, or $44 if paid by F&F.
Thank you.
what type of resistor/capacitor to use in the Zobel network? (LM3886 schematic inside)
- By mrjayviper
- Chip Amps
- 11 Replies
For the capacitor, I have both 0.1uF in ceramic and polypro. Can I use either of them? If yes, which one is preferrable?
Thanks in advance.
Thanks in advance.
how to properly integrate a gain stage in my class a amplifier?
- By No ideas
- Solid State
- 6 Replies
Recently i made a simple common emitter stage on a breadboard that was an altered version of this amplifier from "John audio tech" with and without the constant current source,
the version with the constant current source made around 1.1W of clean power into 8 ohms, and the version without the constant current source made around 0.45W.
The alternations included an additional constant current source version, diffrent resistor values and an single mje3055 npn instead of an darlington, i was really suprised and wanted to expand on this idea and make it higher power and more practical but i soon realised that the bjt common emitter stage is pretty limited and inpractical for anyting highrt power so i made this but now im having trouble with integrating a gain stage into this circuit.
the version with the constant current source made around 1.1W of clean power into 8 ohms, and the version without the constant current source made around 0.45W.
The alternations included an additional constant current source version, diffrent resistor values and an single mje3055 npn instead of an darlington, i was really suprised and wanted to expand on this idea and make it higher power and more practical but i soon realised that the bjt common emitter stage is pretty limited and inpractical for anyting highrt power so i made this but now im having trouble with integrating a gain stage into this circuit.
Variable Q high pass filter schematic?
- By net-david
- Analog Line Level
- 8 Replies
I'm looking for a schematic for a second order high pass filter with the ability to vary the Q from say 0.707 to 2.8 or so. Playing around with an update to the JBL BX63/BX63A. I know I can do it with rotary switches and fixed resistors, but I'd prefer to use a dual gang potentiometer.
From my calculations with C1=C2= 0.22uF, I'd need R1 to vary from 20k to 5.1k while R2 varies from 39k to 150k for Qs from 0.707 to 2.75.
I've searched and not found anything, maybe I'm not using the correct keywords?
From my calculations with C1=C2= 0.22uF, I'd need R1 to vary from 20k to 5.1k while R2 varies from 39k to 150k for Qs from 0.707 to 2.75.
I've searched and not found anything, maybe I'm not using the correct keywords?
For Sale New Old Stock Tektronix P6112 100mHz 10X Oscilloscope Probe
SOLD
New in bag Tektronix P6112 100mHz, 10x. Proper probe for TDS 210 or TDS 220.
Price includes shipping only to CONUS.
$25 if using PP G&S, $24 if using F&F.
Thank you.
New in bag Tektronix P6112 100mHz, 10x. Proper probe for TDS 210 or TDS 220.
Price includes shipping only to CONUS.
$25 if using PP G&S, $24 if using F&F.
Thank you.
Measured woofer response is way off from calculated response.
- Software Tools
- 17 Replies
Greetings. I just carefully designed and built a beautiful set of speakers, but when I measure the woofer low-end response, it's very far from the calculated response. Here are the details.
I'm confused and frustrated. Any advice would be helpful.
View attachment 1358224
- I'm using pairs of CSS WR125 drivers. I got their T/S parameters from the pdf file included in the documents. (see CSS-FR125S-wr125-dimensions)
- I enter these parameters into VituixCAD. and modeled a closed box with Qtc of 0.8. It said 18.6 liters for 2 drivers in parallel. (see VituixCAD showing 18.6 liters)
- So I build a cabinet with 18.1 liters and stuff it with the recommended amount of Polyfill - 1lb/ft^3 I think. I mount the 2 drivers in parallel.
- I made close-mic (nearfield) measurements with the Arta software. I chose one of the two woofers, but kept them both in parallel. I've got a calibrated microphone from Dayton Audio, a nice external power supply with synchronized clocks for the input and output, a nice LM3886 amplifier etc . . Not sure of where to put the microphone, I made several measurements of one woofer from 0.5cm to 5.0 cm from the phase plug, as well as an off-axis measurement. They were all very similar after normalizing absolute SPL levels, so microphone placement is not an issue.
- The FRD data shows a big peak above 100Hz and a 14dB/octave slope!
(see VituixCAD FRD) - I also measure the impedance (of both drivers in parallel) with the Dayton DATS V3. The result says my Qtc=1.18! (see snip of DATS data)
I'm confused and frustrated. Any advice would be helpful.
View attachment 1358224
Attachments
Crossover advice for TAD TL-1601
Hi all,
Measuring TAD TL-1601, I found its frequency response missing 100Hz to 300Hz range. it has a very deep "abyss" in that range (~15dB).
Pls check this pic.
I need advice on how to cross this bass driver, i have tried to cross it with third-order, fourth order crossover (500Hz) but that "abyss" is still there.
Thanks all.
Measuring TAD TL-1601, I found its frequency response missing 100Hz to 300Hz range. it has a very deep "abyss" in that range (~15dB).
Pls check this pic.
I need advice on how to cross this bass driver, i have tried to cross it with third-order, fourth order crossover (500Hz) but that "abyss" is still there.
Thanks all.