Id is dependent on Vgs. You want to have as low gatecharge as possible to reduce the gatecharge before the draincurrent starts to rise. A high impedance driver will raise the Turn off and on time.
That is the reason why the driverstage should be able to sink or source as much current as possible.
I think ... I think some times the problem with mosfet and IGBT outputstage (The sound quality) is addressed to the gatecharge.
If your driverstage cant remove the gatecharge fast enough you will a short shortcut in the outputstage when one mosfet turns off and another turns on wich again can be the reason why MOSFET class A sounds a lot better than MOSFET class AB.
The size off gatecharge is dependent of Vgs and Vds. It would desirable go get as low Vgs and Vds chage as possible.
I dont know if i am right? Ask Nelson ... He does nothing but designing with mosfet... Or ask Elso.
Sonny
That is the reason why the driverstage should be able to sink or source as much current as possible.
I think ... I think some times the problem with mosfet and IGBT outputstage (The sound quality) is addressed to the gatecharge.
If your driverstage cant remove the gatecharge fast enough you will a short shortcut in the outputstage when one mosfet turns off and another turns on wich again can be the reason why MOSFET class A sounds a lot better than MOSFET class AB.
The size off gatecharge is dependent of Vgs and Vds. It would desirable go get as low Vgs and Vds chage as possible.
I dont know if i am right? Ask Nelson ... He does nothing but designing with mosfet... Or ask Elso.
Sonny
Have i understand it right?
You draw a "parallel" line between "Vgs versus gatecharge" and "Vgs versus Id"?
Yes i can You are interrested in running the mosfet at a high Vgs as possible. The Drain current will be high. If you make a change in Id in the order of 2%. Then the Vgs will only change a small amount wich again will result in a small change in gatecharge?
The perfect would then be a mosfet with really high "gfs" to get the Vgs change as small as possible?
Sonny
You draw a "parallel" line between "Vgs versus gatecharge" and "Vgs versus Id"?
Yes i can You are interrested in running the mosfet at a high Vgs as possible. The Drain current will be high. If you make a change in Id in the order of 2%. Then the Vgs will only change a small amount wich again will result in a small change in gatecharge?
The perfect would then be a mosfet with really high "gfs" to get the Vgs change as small as possible?
Sonny
P.Lacombe,
I've checked it out:
there are igbt's made for audio. There are also igbt-amp designs on the net, like:
http://www.arky.ru/audio/shem/igbt/igbt.htm
http://digilander.iol.it/essentialaudio/unetto_home.htm
I've seen a lot of power-mosfets having worser characteristics than those of the power igbts. But you can find good types of mosfets or igbts for audio performance.
Best regards,
HB.
I've checked it out:
there are igbt's made for audio. There are also igbt-amp designs on the net, like:
http://www.arky.ru/audio/shem/igbt/igbt.htm
http://digilander.iol.it/essentialaudio/unetto_home.htm
I've seen a lot of power-mosfets having worser characteristics than those of the power igbts. But you can find good types of mosfets or igbts for audio performance.
Best regards,
HB.
The first link is a design from elektor some years ago. They use Gt20D101 and D201 but the amp was in the begining made to work with MOSFET. This Amp do have higher power rating thoug.
The UNETTO amp uses the same IGBT.
One question i have on my mind. Why do they depent their frequency compensation (miller cap) in the UNE TTO on Q7 or Q8 b-c capacitance!?!?!? You will not get two amps with the same Phasemargin (read sound)!!!!!!!!!
Sonny
The UNETTO amp uses the same IGBT.
One question i have on my mind. Why do they depent their frequency compensation (miller cap) in the UNE TTO on Q7 or Q8 b-c capacitance!?!?!? You will not get two amps with the same Phasemargin (read sound)!!!!!!!!!
Sonny
Thanks for looking up the Id/Qgs for me, Nelson. 
This is interesting - in the "active region" the drain current is approximately linear to the integral of gate current.
Some more thoughts on the use of FETs:
I notice quite often designers treating FETs like substitutes for BJTs, especially in emitter-follower output stages. This really doesn't make much sense to me. Using a FET as a source-follower is undermined because of the transconductance characteristic and the very high Cgs and Cdg. So they don't work as well as BJTs as emitter-follower buffers. However, they could be used as current amps as part of a BJT emitter-follower output. Using power FETs in common-source is more likely to be fruitful. Again, remember the Cdg which is usually very large (>200pF) and varies non-linearly and inversely with Vdg. This can cause some interesting "Miller" effects if not dealt with. And the transconductance is non-linear too. Uncompensated, these characteristics will degrade the sound quality, especially so if loop feedback is used (yes, feedback can make the sound much worse and/or cause instability). Another pointer is to not assume a mfrs complementary device is truely complementary - with some power BJTs they are pretty darned close but power FETs tend to be miles apart on capacitances and transconductance curves, so look for the best complement not just the one with the same serial no preceeded by a "9". Of course, if you choose a single-ended stage you don't have to worry about this - instead you have to worry about heat.
This is interesting - in the "active region" the drain current is approximately linear to the integral of gate current.Some more thoughts on the use of FETs:
I notice quite often designers treating FETs like substitutes for BJTs, especially in emitter-follower output stages. This really doesn't make much sense to me. Using a FET as a source-follower is undermined because of the transconductance characteristic and the very high Cgs and Cdg. So they don't work as well as BJTs as emitter-follower buffers. However, they could be used as current amps as part of a BJT emitter-follower output. Using power FETs in common-source is more likely to be fruitful. Again, remember the Cdg which is usually very large (>200pF) and varies non-linearly and inversely with Vdg. This can cause some interesting "Miller" effects if not dealt with. And the transconductance is non-linear too. Uncompensated, these characteristics will degrade the sound quality, especially so if loop feedback is used (yes, feedback can make the sound much worse and/or cause instability). Another pointer is to not assume a mfrs complementary device is truely complementary - with some power BJTs they are pretty darned close but power FETs tend to be miles apart on capacitances and transconductance curves, so look for the best complement not just the one with the same serial no preceeded by a "9". Of course, if you choose a single-ended stage you don't have to worry about this - instead you have to worry about heat.
Nelson Pass said:
Nelson,
Thank you for your reply!
I really don't have any way to measure distortion except to put the amp on an oscilloscope and do an X-Y plot. I built a Zen using an IRF130 a couple of years ago. (because that device was on hand) A few months I built my present amp using the 1058's which is actually two Zen amps bridged to eliminate DC across the speaker with using a blocking capacitor. (and fed push-pull) To me the newer amp sounds better, but there is more than one variable here (lack of output cap - which was a cheapo I admit, less voltage swing per device, etc.) and of course I am not without bias. I still have the single ended Zen and will put the new and old on the 'scope in the next day or two to see if I can see any difference.
Thanks again for your time and all you have done for us DIYers!
I would be willing to bet that with a single amp
you'll see more measured distortion. I have seen
and built Zens with the lower transconductance devices
and they all measured greater distortion, and to my ear,
did not sound as good.
One of the trade-offs is the lower capacitance, so that
with lower transcondance devices you can use a higher
input impedance, and this sometimes helps with sources
that have trouble driving the low input impedance.
If you are driving the speaker with 2 balanced Zens
(or better yet 2 pairs of parallel balanced Zens) you
can get considerably less distortion due to cancellation
of the dominant second harmonic.
you'll see more measured distortion. I have seen
and built Zens with the lower transconductance devices
and they all measured greater distortion, and to my ear,
did not sound as good.
One of the trade-offs is the lower capacitance, so that
with lower transcondance devices you can use a higher
input impedance, and this sometimes helps with sources
that have trouble driving the low input impedance.
If you are driving the speaker with 2 balanced Zens
(or better yet 2 pairs of parallel balanced Zens) you
can get considerably less distortion due to cancellation
of the dominant second harmonic.
Circlotron:
>is there such a thing as a power JFET?<
At least there used to be, from the likes of Yamaha, Sony and NEC. These devices were referred to as vertical power FETs, and date back to the mid-70s, right around the time that Hitachi was introducing its MOSFETs (which originally had a completely different designation than 2SK134/2SJ48 et al). As far as I know, there were a number of application issues relating primarily to the gate drive which made these vertical power FETs tricky to use and easy to break.
In most cases, the devices were featured in commercial power amps from their respective manufacturers, and I would not be surprised if the audio divisions in each company had a major influence in getting these devices produced.
A quick list of commercial power amps incorporating these devices would include JVC's JM-S7, Sony's TA-5650 and TAN-5550, Sansui's BA-1000, and of course the B-1 from Yamaha. I am sure that there are many more that I have forgotten.
A few years ago, Tokin started manufacturing a line of static-induction transistors (SIT), which behave very similarly to the older Sony, Yamaha and NEC devices. However, I think that Tokin was subsequently acquired by NEC, and I haven't kept track of what happened to the SIT lineup.
2SK60/2SJ18 (Sony), 2SK70/2SJ20 (NEC), and 2SK77 (Yamaha) are some of the device codes that I remember. Apologies for any memory lapses.
regards, jonathan carr
>is there such a thing as a power JFET?<
At least there used to be, from the likes of Yamaha, Sony and NEC. These devices were referred to as vertical power FETs, and date back to the mid-70s, right around the time that Hitachi was introducing its MOSFETs (which originally had a completely different designation than 2SK134/2SJ48 et al). As far as I know, there were a number of application issues relating primarily to the gate drive which made these vertical power FETs tricky to use and easy to break.
In most cases, the devices were featured in commercial power amps from their respective manufacturers, and I would not be surprised if the audio divisions in each company had a major influence in getting these devices produced.
A quick list of commercial power amps incorporating these devices would include JVC's JM-S7, Sony's TA-5650 and TAN-5550, Sansui's BA-1000, and of course the B-1 from Yamaha. I am sure that there are many more that I have forgotten.
A few years ago, Tokin started manufacturing a line of static-induction transistors (SIT), which behave very similarly to the older Sony, Yamaha and NEC devices. However, I think that Tokin was subsequently acquired by NEC, and I haven't kept track of what happened to the SIT lineup.
2SK60/2SJ18 (Sony), 2SK70/2SJ20 (NEC), and 2SK77 (Yamaha) are some of the device codes that I remember. Apologies for any memory lapses.
regards, jonathan carr
Apparently everyone here is convinced that power bipolar junction transistor device capacitances are smaller than power MOSFET device capacitances. This may be true when you compare the bipolar part's junction capacitances to the MOSFET's gate and overlap capacitances, but as soon as you forward bias a bipolar device, you get a large diffusion capacitance between base and emitter (this capacitance models the minority charge storage in the base). For a typical epitaxial-base power transistor with 10MHz fT biased at 1A, its value is around 600nF, much greater than the oxide capacitance of a normal power MOSFET. In fact, this is why the fT of the transistor is only 10MHz despite of its huge transconductance.
Interesting argument, Marcel. You may be right about the base capacitance being very large: how did you arrive at these figures?
One thing that may be of more importance is the energy required to cause a change in output current. For example, for a FET and BJT with similar Pd and Imax, how much charge is required to change their output currents from, say, 1A to 2A in 1us? You would have to make some assumption about the change in collector-base/drain-gate voltage to take account of charging Ccb/Cdg.
Figuring out the gate charge is relatively easy because the datasheets usually show a graph of Id vs Q as FETs are most often used in switching applications. Trickier to find the equivalent for a BJT, perhaps.
One thing that may be of more importance is the energy required to cause a change in output current. For example, for a FET and BJT with similar Pd and Imax, how much charge is required to change their output currents from, say, 1A to 2A in 1us? You would have to make some assumption about the change in collector-base/drain-gate voltage to take account of charging Ccb/Cdg.
Figuring out the gate charge is relatively easy because the datasheets usually show a graph of Id vs Q as FETs are most often used in switching applications. Trickier to find the equivalent for a BJT, perhaps.
Marcel,
Accepted, but you refer only to base/emitter capacitance, which is not too significant, particularly in a common collector since the Vbe does not much change during operation.
Of far greater significance is the depletion capacitance across the base/collector, the so-called Miller capacitance, which is profoundly influential in the common emitter configuration.
Your comment is interesting, but less relevant in this context, since the gate capacitance of a mosfet is far higher wrt the drain, and this effectively mandates use of muscular drive, not so much a problem with a bipolar output stage and in any case ameliorated by use of a double emitter follower.....
In closing I'd suggest the propensity to self-oscillation of the mosfet introduces other problems, necessitating a gate stopper which to some extent negates the use of strong drive at the gate.
I apologize for the subjective comments; I don't have capacitance figures to hand, but am flying blind. I do know the gate capacitance of a P type IRF9140 is 600pF, and this capacitance is against the drain, a real PITA for a source follower.
Cheers,
Hugh
Accepted, but you refer only to base/emitter capacitance, which is not too significant, particularly in a common collector since the Vbe does not much change during operation.
Of far greater significance is the depletion capacitance across the base/collector, the so-called Miller capacitance, which is profoundly influential in the common emitter configuration.
Your comment is interesting, but less relevant in this context, since the gate capacitance of a mosfet is far higher wrt the drain, and this effectively mandates use of muscular drive, not so much a problem with a bipolar output stage and in any case ameliorated by use of a double emitter follower.....
In closing I'd suggest the propensity to self-oscillation of the mosfet introduces other problems, necessitating a gate stopper which to some extent negates the use of strong drive at the gate.
I apologize for the subjective comments; I don't have capacitance figures to hand, but am flying blind. I do know the gate capacitance of a P type IRF9140 is 600pF, and this capacitance is against the drain, a real PITA for a source follower.
Cheers,
Hugh
Regarding Traderbam's question: in the following I will assume you bias your transistor in the normal forward active region, not in saturation or in reverse or anything funny. When you know a transistor's transconductance and fT for a given bias point, the sum of the base-emitter and base-collector capacitances is given by the equation:
cpi+cmu~=gm/(2*pi*fT),
where cpi is the base-emitter capacitance, cmu is the base-collector capacitance, gm is the transconductance, fT is the extrapolated frequency at which the current gain drops to unity and pi is 3.14159265358979....
Ideally, neglecting some second-order effects which occur at high current densities, the transconductance of a bipolar transistor equals:
gm=IC*q/(kT),
where IC is the collector bias current, q is the electron charge (1.6022E-19 C), k Boltzmann's constant (1.38065E-23 J/K) and T is the absolute temperature.
Assuming 10MHz fT, 1A bias current and kT/q=26mV (which is true just above room temperature):
gm~=38.4615 S,
cpi+cmu~=612.134nF.
The collector-base capacitance cmu is just pure junction capacitance, the base-emitter capacitance cpi is a combination of junction and diffusion capacitance. So after subtracting the junction capacitances, you are left with the base-emitter diffusion capacitance.
Regarding Hugh's/AKSA's remarks, it is certainly true that capacitance between base and collector is much worse than an equal capacitance between base and emitter. This does, however, not mean that diffusion capacitance is necessarily negligible.
For example, without any cascoding, base-collector capacitance is gm*Rload+1 times as bad as base-emitter capacitance, due to the Miller effect in common-emitter stages or the bootstrapping effect in emitter followers. When gm=38 siemens and the load resistance Rload=8 ohm, gm*Rload+1=305. But a 600nF cpi then still has a comparable influence as a cmu of 600nF/305, or almost 2nF. The situation gets worse at lower load resistances.
To make matters worse, I think that in most practical cases in audio amplifiers the static base current required by bipolar output transistors is even larger than the capacitive currents. Of course you can solve that by using double emitter followers, at the expense of a more complicated high frequency behaviour of the feedback loop, but you could do the same in an amplifier with a MOSFET common-drain output stage as long as gate stoppers don't spoil the fun.
By the way, has anyone ever tried RC damping networks between gate and source instead of gate series resistors to stop parasitic oscillations in power MOSFET's? I mean connecting a resistor in series with a capacitor and then putting the whole thing between gate and source. I haven't tried it, but I have often used circuits like that in unstable RF IC's.
Yours sincerely,
Marcel
cpi+cmu~=gm/(2*pi*fT),
where cpi is the base-emitter capacitance, cmu is the base-collector capacitance, gm is the transconductance, fT is the extrapolated frequency at which the current gain drops to unity and pi is 3.14159265358979....
Ideally, neglecting some second-order effects which occur at high current densities, the transconductance of a bipolar transistor equals:
gm=IC*q/(kT),
where IC is the collector bias current, q is the electron charge (1.6022E-19 C), k Boltzmann's constant (1.38065E-23 J/K) and T is the absolute temperature.
Assuming 10MHz fT, 1A bias current and kT/q=26mV (which is true just above room temperature):
gm~=38.4615 S,
cpi+cmu~=612.134nF.
The collector-base capacitance cmu is just pure junction capacitance, the base-emitter capacitance cpi is a combination of junction and diffusion capacitance. So after subtracting the junction capacitances, you are left with the base-emitter diffusion capacitance.
Regarding Hugh's/AKSA's remarks, it is certainly true that capacitance between base and collector is much worse than an equal capacitance between base and emitter. This does, however, not mean that diffusion capacitance is necessarily negligible.
For example, without any cascoding, base-collector capacitance is gm*Rload+1 times as bad as base-emitter capacitance, due to the Miller effect in common-emitter stages or the bootstrapping effect in emitter followers. When gm=38 siemens and the load resistance Rload=8 ohm, gm*Rload+1=305. But a 600nF cpi then still has a comparable influence as a cmu of 600nF/305, or almost 2nF. The situation gets worse at lower load resistances.
To make matters worse, I think that in most practical cases in audio amplifiers the static base current required by bipolar output transistors is even larger than the capacitive currents. Of course you can solve that by using double emitter followers, at the expense of a more complicated high frequency behaviour of the feedback loop, but you could do the same in an amplifier with a MOSFET common-drain output stage as long as gate stoppers don't spoil the fun.
By the way, has anyone ever tried RC damping networks between gate and source instead of gate series resistors to stop parasitic oscillations in power MOSFET's? I mean connecting a resistor in series with a capacitor and then putting the whole thing between gate and source. I haven't tried it, but I have often used circuits like that in unstable RF IC's.
Yours sincerely,
Marcel
Hi all,
for 10 years I was making amplifiers with N-channel only MOSFETs in the output and really like these devices. Main reason is that the modern "switching" logic level MOSFETS, like BUK555-100 and later similar devices, are incredibly tightly matched inside one batch (less than 1% difference in capacitances, threshold voltage and transconductance). So you have nice symmetry for push-pull without spending much money (BUK555 was about $1 each) . The main advantage of the MOSFET ouput stage, in my view, is that it has low output impedance even without NFB, and does not change the load of the voltage amplifier as much as BJT output does with the load change.
Al
for 10 years I was making amplifiers with N-channel only MOSFETs in the output and really like these devices. Main reason is that the modern "switching" logic level MOSFETS, like BUK555-100 and later similar devices, are incredibly tightly matched inside one batch (less than 1% difference in capacitances, threshold voltage and transconductance). So you have nice symmetry for push-pull without spending much money (BUK555 was about $1 each) . The main advantage of the MOSFET ouput stage, in my view, is that it has low output impedance even without NFB, and does not change the load of the voltage amplifier as much as BJT output does with the load change.
Al
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