Don, all i can say is THANK YOU for this! Without wanting to sound insulting to many here, I am simply not old enough to be that experienced with tubes, especially since there is a large non-practitioning gap with me and tubes between my HS days and perhaps 4-5 years ago - but what you state here, and i have emphasized, is alsmot exactly along the lines of my own thoughts re UL - which, given the H7K paper and the many existing UL constructions past and present, I simply kept to myself, thinking there is probably something I am missing.smoking-amp said:
It has been advocated here before, that pentodes get more linear with lower screen voltages, which would of course seem almost obvious given what G2 does in a pentode. Various 'kinky' troubles begin whenever t he plate goes near or, worse, below G2 potential. In the classical Ul this is the default case at low power. I think a search would reveal several instances where it was said, with tube curves at hand, that G2 should be kept well away in potential from the plate - this is a simple corroboration of what you are saying. I am looking forward to your measurements.
I also agree with your assertion that there is no universally acceptable 'ideal' UL ratio, and that it depends on the combination of tube and transformer, if you model the various feedbacks as a different kind of triode, this follows automatically. There have been things on the web that deal with this, and one of them was based on the electrode distance from cathode, and (supposedly) resulting Rp per unit area of G2 and plate - both, in a sense being 'plate'. i will try to reference a link, can't find it now, but would love to read your comment on this.
EDIT: someone else was typing about it as i was typing my own relply - see post above mine, or go here:
The resource cannot be found.
Hey-Hey!!!,
If we're to believe the curves, the plate can go a fair distance below g2 before the screen current begins to climb. It is dependant on the type of pentode of course, and some are better than others.
I don't see any decrease in linearity as the plate goes below g2 either. If the load line is reasonable drawn, the 'kinky' behaviour isn't anywhere the load line. Now if one accounts for an inductive elipse, the line begins to blur(so to speak ).
cheers,
Douglas
Hi ilimzn, and isaacc7,
Thanks for the encouragement, I will continue to plow on along these lines.
I read the Grimwood Optimised UL article last night as Johan cited this same article for a UL characteristic chart for the KT88. It seems to be a very LONG winded suggestion that the DC fields in the tube should be partitioned in the same way as the AC fields are naturally due to element spacing or geometry. This seems fine as far as avoiding tube voltage breakdown problems, but this will certainly greatly limit the maximum power that can be derived from it's operation. But I think his emphasis is mainly on the quality or linearity of amplification and avoidance of instabilities, particlularly for difficult tubes. He could be right, I have no idea. Some spectrum/distortion tests would seem to be called for to resolve this hypothesis. Grimwood does not provide this (well I didn't read the other UL pages he referenced, maybe he does?), but depends on listening results apparently. I would say this would be a good little test project for someone interested, just need a sound card.
Anyway, I woke up this morning and realized that I have been blab.. blab ...blab.. about my new effective triode formulas, but haven't applied them to H-K's article to see if any insight could be gained about their magic 43% tap. so here goes:
They mention that the power supply current for their circuit reaches 130 mA peak current. Looking into the GE databook for the 6L6 (for two tubes P-P in class A), they conveniently list 134 mA operation for two tubes as giving a gm1 of 5700. The triode wired Mu for the 6L6 is listed as 8. Since Mu = gm1 * Rp for triodes, this would mean the triode wired 6L6 config has an Rp of 1403 OHMs.
Next, they hint that the idle two tube current is 100mA as can be seen from the schematic with 35V across the 350 OHM common cathode resistor. This is clearly an amplifier that is mostly operating in class A operation, if not fully. Even for class A, the power supply draw increases a little at peak power due to the residual 3/2 power law nonlinearity of the loaded output tubes.
Next, they use a 6600 OHM CT output transformer. Which would give 1/4 that Z for each tube in class B but 1/2 that Z in class A (in class A, both tubes are operating simulataneously, so each tube provides 1/2 the current for the observed voltage swing, so each sees the load as 2x)
Using the class A operating mode we then get 3300 OHM seen by each tube.
Now let us apply the effective triode formula for Rp and Mu using UL% at .43 (the winding tap is at 43%, their chart gives impedance ratios on the X axis which is .43 * .43 = .185 or 18.5%)
effective Rp = triode_Rp/ .43
so eff. Rp = 1403/ .43 => 3263 OHM
and eff. Mu => 8/.43 => 18.6 (if we had the grid1 drive signal level, we could compute the maximum power output using this)
3263 OHm eff. Rp sure looks like a close match to 3300 OHM loading Z, this is the condition for maximum power transfer from a triode when load = Rp. I think we have seen the smoking gun here!!
We can now likely compute the optimum %UL for other tubes (with different triode MU and GM1 and Gm2) used with any Z primary transformer. So I think this 43% magic tap is just an artifact of using a 6L6 with a 6600 OHM P-P transformer.
Now if H-K had just done this analysis, I would not be "%*/!^*" on their paper today.
Don
Thanks for the encouragement, I will continue to plow on along these lines.
I read the Grimwood Optimised UL article last night as Johan cited this same article for a UL characteristic chart for the KT88. It seems to be a very LONG winded suggestion that the DC fields in the tube should be partitioned in the same way as the AC fields are naturally due to element spacing or geometry. This seems fine as far as avoiding tube voltage breakdown problems, but this will certainly greatly limit the maximum power that can be derived from it's operation. But I think his emphasis is mainly on the quality or linearity of amplification and avoidance of instabilities, particlularly for difficult tubes. He could be right, I have no idea. Some spectrum/distortion tests would seem to be called for to resolve this hypothesis. Grimwood does not provide this (well I didn't read the other UL pages he referenced, maybe he does?), but depends on listening results apparently. I would say this would be a good little test project for someone interested, just need a sound card.
Anyway, I woke up this morning and realized that I have been blab.. blab ...blab.. about my new effective triode formulas, but haven't applied them to H-K's article to see if any insight could be gained about their magic 43% tap. so here goes:
They mention that the power supply current for their circuit reaches 130 mA peak current. Looking into the GE databook for the 6L6 (for two tubes P-P in class A), they conveniently list 134 mA operation for two tubes as giving a gm1 of 5700. The triode wired Mu for the 6L6 is listed as 8. Since Mu = gm1 * Rp for triodes, this would mean the triode wired 6L6 config has an Rp of 1403 OHMs.
Next, they hint that the idle two tube current is 100mA as can be seen from the schematic with 35V across the 350 OHM common cathode resistor. This is clearly an amplifier that is mostly operating in class A operation, if not fully. Even for class A, the power supply draw increases a little at peak power due to the residual 3/2 power law nonlinearity of the loaded output tubes.
Next, they use a 6600 OHM CT output transformer. Which would give 1/4 that Z for each tube in class B but 1/2 that Z in class A (in class A, both tubes are operating simulataneously, so each tube provides 1/2 the current for the observed voltage swing, so each sees the load as 2x)
Using the class A operating mode we then get 3300 OHM seen by each tube.
Now let us apply the effective triode formula for Rp and Mu using UL% at .43 (the winding tap is at 43%, their chart gives impedance ratios on the X axis which is .43 * .43 = .185 or 18.5%)
effective Rp = triode_Rp/ .43
so eff. Rp = 1403/ .43 => 3263 OHM
and eff. Mu => 8/.43 => 18.6 (if we had the grid1 drive signal level, we could compute the maximum power output using this)
3263 OHm eff. Rp sure looks like a close match to 3300 OHM loading Z, this is the condition for maximum power transfer from a triode when load = Rp. I think we have seen the smoking gun here!!
We can now likely compute the optimum %UL for other tubes (with different triode MU and GM1 and Gm2) used with any Z primary transformer. So I think this 43% magic tap is just an artifact of using a 6L6 with a 6600 OHM P-P transformer.
Now if H-K had just done this analysis, I would not be "%*/!^*" on their paper today.
Don
Of course, you are right, i was generalizing too much... after all, heaps of work have been done on this phenomenon, resulting in, amongst other, shadow grid tubes and as a subset, beam tetrodes. but i am sure you understand the gist of what I was saying - in the classical UL with DC G2 and plate voltages the same, i.e. just an UL tap on the output transforme, and not a separate winding, whenever the plate swings under the supply, it is going below G2 potential - and above G2 potential when it is swinging over the supply. The fact that UL tap is by definition <100% makes it so.
In general, pentodes and beam tetrodes tend to be less kinky over the useful plate voltage range (just look at sweep tubes!), the lower the G2 voltage - but of course, you sacrifice gm by lowering Vg2. In an implementation where G2 is driven via a separate winding with it's own G2 supply, it follows that one could optimize sensitivity vs distortion for a given percentage UL, by chosing the right DC Vg2, which would alow the designer to move the point at which the plate swings below G2 and G2 starts capturing more current, in esence, pushing as much of any kink to higher swings. This is particulairly important with your excellent point about drawing a complex load ellipse - in such an opimized system, there would be less 'blur' (good description, BTW).
smoking-amp said:And... we would not be stuck with all these 43% tapped output transformers on the market that are totally meaningless!
Don
So get yer winders to build at something other than 43%. IIRC, from looking at a test of output TX's few had a tap at 43%. 25-50% was the published spread...
I read one of H&K's papers on the taping point, and they said that for the 6V6 22.5% was the sweet spot. Now the Z565 of theirs came in at 25% taping.There are of course limits on the taping, as end-of-layer points are the most convenient places to tap. Get several! The last pair I ordered of the Peerless S265Q came with 3. Being able to hook up the g2 at another spot is quite useful, and my circuits sometimes use more than one pair of taps anyway...
cheers,
Douglas
"Care to explain the EL84 datasheet? There certainly seems to be something good happening at 43% (as opposed to 20%)."
OK. I assume you mean the Mullard data sheets. Using page C14 for 43% with 6000 OHM CT P-P xfmr. For the 10Watt output max shown, Ik is at 94 mA. I am assuming class A operation here (the GE data book shows 48 mA idle per tube for class A) so at peak output, one tube would be at this 94mA peak and the other at 0 mA. For 94 mA, page C6 of the Mullard sheets shows:
Ra = 1360 Ohm
Mu_triode = 21.9
Gm = 16.1 ma/V
Using the effective triode formula: eff. Rp = Ra/.43 => 3162 OHM
The 6000 OHM P-P CT xfmr in class A would be loading at 3000 OHM per tube. This is a pretty good match. 45% UL would be perfect.
Mu effective would be: Mu_triode/.43 => 50.9
Vin on the Mullard chart C14 is 16.5 V g-g rms => 11.66 Vpk on each grid.
Output voltage swing then would be 11.66 *50.9 => 593 Vpk for unloaded or 593/2 for matched Z load (Rp = Zload) => 296.5 Vpk which matches "roughly" with the 300V B+ available.
Mu_triode on chart C6 is actually varying between 14 and 21.9 between 0 and 94 mA, so if we take the average of 17.95 for Mu_triode, and times 1/.43 to get the effective triode average MU:
we get 11.66*17.95/.43 => 486.7 Vpk unloaded and 486.7/2 => 243 Vpk loaded (Rp = Zload) which is more plausible with a 300V B+.
Calculating power out then: 243 *243 / 3000 = 19.68 W pk power or 9.8 Watt average. Not a bad match to the 10W shown on the graph.
Seems to me the effective triode #s hold up.
The EL84 is not much different in Rp from the 6L6, (1360 vs 1403 Ohm) so not too surprising to get near 43% again with a 6K xfmr.
The 20% UL case would like to see more like a 7K or 8K OHm loading (per tube) which would require a 16K Ohm P-P Ct xfmr (in class A, 32K Ohm in class B). So the Mullard charts for 20% are grossly overloading the tube. Hence the high distortion.
Anyone have data for UL with a very different tube?
Don
OK. I assume you mean the Mullard data sheets. Using page C14 for 43% with 6000 OHM CT P-P xfmr. For the 10Watt output max shown, Ik is at 94 mA. I am assuming class A operation here (the GE data book shows 48 mA idle per tube for class A) so at peak output, one tube would be at this 94mA peak and the other at 0 mA. For 94 mA, page C6 of the Mullard sheets shows:
Ra = 1360 Ohm
Mu_triode = 21.9
Gm = 16.1 ma/V
Using the effective triode formula: eff. Rp = Ra/.43 => 3162 OHM
The 6000 OHM P-P CT xfmr in class A would be loading at 3000 OHM per tube. This is a pretty good match. 45% UL would be perfect.
Mu effective would be: Mu_triode/.43 => 50.9
Vin on the Mullard chart C14 is 16.5 V g-g rms => 11.66 Vpk on each grid.
Output voltage swing then would be 11.66 *50.9 => 593 Vpk for unloaded or 593/2 for matched Z load (Rp = Zload) => 296.5 Vpk which matches "roughly" with the 300V B+ available.
Mu_triode on chart C6 is actually varying between 14 and 21.9 between 0 and 94 mA, so if we take the average of 17.95 for Mu_triode, and times 1/.43 to get the effective triode average MU:
we get 11.66*17.95/.43 => 486.7 Vpk unloaded and 486.7/2 => 243 Vpk loaded (Rp = Zload) which is more plausible with a 300V B+.
Calculating power out then: 243 *243 / 3000 = 19.68 W pk power or 9.8 Watt average. Not a bad match to the 10W shown on the graph.
Seems to me the effective triode #s hold up.
The EL84 is not much different in Rp from the 6L6, (1360 vs 1403 Ohm) so not too surprising to get near 43% again with a 6K xfmr.
The 20% UL case would like to see more like a 7K or 8K OHm loading (per tube) which would require a 16K Ohm P-P Ct xfmr (in class A, 32K Ohm in class B). So the Mullard charts for 20% are grossly overloading the tube. Hence the high distortion.
Anyone have data for UL with a very different tube?
Don
"So get yer winders to build at something other than 43%."
Well, I did special order the Edcor CXPP100-SP1 output xfmr with 20% separate windings. Was more for CFB experimenting, but now I'll have to find a tube to try 20% UL too.
I should point out however, that this "optimum" %UL for power is just making Zload = Rp effective. This gives maximum power out, but not necessarily lowest distortion. The H-K UL data is really "rigged" in my opinion since they test at the maximum power out drive for all cases.
This causes serious overdrive or clipping for the other "non optimum" %UL in their test. Making everything else not at the "optimum %UL" look horrible distortion-wise in their test.
A more sane test would be to test at below clipping and G1 current overdrive and just measure clean power out and distortion.
Almost certainly, lower distortion than at the power "optimum %UL" will result with Zloading > than Rp effective, just like any normal triode design.
Don
Well, I did special order the Edcor CXPP100-SP1 output xfmr with 20% separate windings. Was more for CFB experimenting, but now I'll have to find a tube to try 20% UL too.
I should point out however, that this "optimum" %UL for power is just making Zload = Rp effective. This gives maximum power out, but not necessarily lowest distortion. The H-K UL data is really "rigged" in my opinion since they test at the maximum power out drive for all cases.
This causes serious overdrive or clipping for the other "non optimum" %UL in their test. Making everything else not at the "optimum %UL" look horrible distortion-wise in their test.
A more sane test would be to test at below clipping and G1 current overdrive and just measure clean power out and distortion.
Almost certainly, lower distortion than at the power "optimum %UL" will result with Zloading > than Rp effective, just like any normal triode design.
Don
Yipes,
When I post on my lonesome own everybody else is sleeping, and as soon as I go to sleep everybody else posts, quietly move threads elsewhere, etc. [oh, and Smoking-amp is introducing new maths by dividing by 0. At 0% tap the rp of a pentode is not infinity! ]
But there are still 3 things that perplexes me (OK, now I must use There) I see the webace article has been posted 3 times, but with little cognisance or at least mention of what the GE characteristics for the KT88 tell us.
Sorry to sound like a skipping CD, but I thought it clearly shows that impedance is not critically related to all of this subject matter, apart from obviously influencing the max. available output power. Why do I get the impression we are still hunting for an optimum here, at least regarding the UL topology?
Secondly it seems to be clear why 43% is popular as an "optimum" (sic) tap. Maximum output power stays about the same, but rp goes from 8K for 20% G2 taps to 4,1K for 45% taps in the KT88 graphs. Distortion is slightly lower (maybe not worth mentioning, but still). This is by the way very similar to EL34 behaviour (the Mullard graphs are on the internet, as are those for EL84). Ditto for my tests with 6L6, although I use 33% taps - windings come out conveniently that way.
So praythee, what do we have against 43% or thereabouts?? I have not seen different information, and I do not contest the maths of Smoking-amp - neither has he shown that there is a substantial advantage at lower taps (if I did not misread somewhere in the dead of night). I get the impression that many of us are simply "wondering", again as if this is the first time someone has thought of this.
One could notice something else that is different, if not much. The EL34 graphs indicate max. available power output for pentode operation, immediately starting to drop somewhat as the screen is shifted, while for the KT88 (and other beam tubes) there is an increase (for RL=6K) from 47W for pentode mode to 50W for early tapped mode - obviously the contribution of the screen grid to usefull output power. (BTW, it would also seem that we have nothing to worry regarding the first watt, judging from both figures 2 and 3 in the Williamson-Walker article, apart from other references.)
Finally I detect here an unexpected nervousness concerning the undesirability of operating the anode and screen at the same dc voltage.
Hey???
Am I missing something after 50 years in the business? For ever amplifiers have worked under most strenuous conditions in this mode - one is not talking about a few volt drop across a power supply choke - and tubes have lasted for years ... Then there are the manufacturers graphs - a 6L6 say, is operational down to below 100Va with the screen still at all of 400V. (Of course handling an ac signal; the next moment the plate shoots up to 700V - nobody is talking of dc amplifiers). Is somebody missing the fact somewhere that the screen grid is not solid, and can never collect nearly a large percentage of the anode current especially in a beam tube? As I have respectfully asked before, are we challenging manufacturers' data? For most of the regular power output tubes anode and screen current/dissipation are given at zero signal as well as maximum output conditions. The rated screen dissipation takes this into account! Surely nobody can suggest that tube manual compilers have forgotten that sometimes a signal must also be handled by the tube.
I am ready to be informed of where I am missing something vital. But I hope to be forgiven for being sceptical about the sudden questioning of practices that existed successfully for more than a half century. It is really unlikely that errors in basic tube operation will now come to light, that failed to do so in the hey-day of the product.
(I see someone else has posted while I was typing - ah! I'm not alone! - Hope not to conflict or repeat.)
Regards all.
When I post on my lonesome own everybody else is sleeping, and as soon as I go to sleep everybody else posts, quietly move threads elsewhere, etc.
[oh, and Smoking-amp is introducing new maths by dividing by 0. At 0% tap the rp of a pentode is not infinity! ]But there are still 3 things that perplexes me (OK, now I must use
There) I see the webace article has been posted 3 times, but with little cognisance or at least mention of what the GE characteristics for the KT88 tell us. Sorry to sound like a skipping CD, but I thought it clearly shows that impedance is not critically related to all of this subject matter, apart from obviously influencing the max. available output power. Why do I get the impression we are still hunting for an optimum here, at least regarding the UL topology?
Secondly it seems to be clear why 43% is popular as an "optimum" (sic) tap. Maximum output power stays about the same, but rp goes from 8K for 20% G2 taps to 4,1K for 45% taps in the KT88 graphs. Distortion is slightly lower (maybe not worth mentioning, but still). This is by the way very similar to EL34 behaviour (the Mullard graphs are on the internet, as are those for EL84). Ditto for my tests with 6L6, although I use 33% taps - windings come out conveniently that way.
So praythee, what do we have against 43% or thereabouts?? I have not seen different information, and I do not contest the maths of Smoking-amp - neither has he shown that there is a substantial advantage at lower taps (if I did not misread somewhere in the dead of night). I get the impression that many of us are simply "wondering", again as if this is the first time someone has thought of this.
One could notice something else that is different, if not much. The EL34 graphs indicate max. available power output for pentode operation, immediately starting to drop somewhat as the screen is shifted, while for the KT88 (and other beam tubes) there is an increase (for RL=6K) from 47W for pentode mode to 50W for early tapped mode - obviously the contribution of the screen grid to usefull output power. (BTW, it would also seem that we have nothing to worry regarding the first watt, judging from both figures 2 and 3 in the Williamson-Walker article, apart from other references.)
Finally I detect here an unexpected nervousness concerning the undesirability of operating the anode and screen at the same dc voltage.
Hey???
Am I missing something after 50 years in the business? For ever amplifiers have worked under most strenuous conditions in this mode - one is not talking about a few volt drop across a power supply choke - and tubes have lasted for years ... Then there are the manufacturers graphs - a 6L6 say, is operational down to below 100Va with the screen still at all of 400V. (Of course handling an ac signal; the next moment the plate shoots up to 700V - nobody is talking of dc amplifiers). Is somebody missing the fact somewhere that the screen grid is not solid, and can never collect nearly a large percentage of the anode current especially in a beam tube? As I have respectfully asked before, are we challenging manufacturers' data? For most of the regular power output tubes anode and screen current/dissipation are given at zero signal as well as maximum output conditions. The rated screen dissipation takes this into account! Surely nobody can suggest that tube manual compilers have forgotten that sometimes a signal must also be handled by the tube.
I am ready to be informed of where I am missing something vital. But I hope to be forgiven for being sceptical about the sudden questioning of practices that existed successfully for more than a half century. It is really unlikely that errors in basic tube operation will now come to light, that failed to do so in the hey-day of the product.
(I see someone else has posted while I was typing - ah! I'm not alone! - Hope not to conflict or repeat.)
Regards all.
Hi Johan and all,
Firstly, the full effective triode formulas do not go to infinity at %UL = 0, I just said that the plate term in the formula was small so it could be ignored for most cases and the formulas simplified to 1/UL% form. With the plate term included, the Rp does indeed converge on the pentode Rp and the Mu effective does converge on the pentode effective Mu. Here are the actual formulas:
Mu_eff = -dVp/dVg = 1/ [ (U%/Mu_scrn) + (1/Mu_plate) ]
Rp_eff = dVp/dI = [K*Ik^(-1/3)] / [ (U%/Mu_scrn) + (1/Mu_plate) ]
or: Rp_eff = K * Mu_eff *Ik^(-1/3)
and Gm = Ik^(1/3) / K ( K is a constant for units and tube geometry, Ik is cathode current )
which nicely gives: Mu_eff = Gm * Rp_eff ( just like any triode! )
OK, next on the KT88 curves. The tube Rp is affected by the plate (or cathode) current (see formula above, Rp proportional to Ik^-1/3). These KT88 curves are scaled power-wise so that the same Rp (or Zout on the graph) occurs for each case, hence the inclusion of only one Zout curve on the graph. Due to this, one expects all the curves to peak at the same %UL. This does not tell us much. The info we want has been factored out.
Now it could be that the power curves shown are the maximum power that can be achieved at some max tube dissipation, and then we could conclude that Ik^1/3 conspires to keep the peak at nearly the same %UL for all cases of xfmr Zload for this tube.
But this favored %UL will still be specific to the particular tube chosen because it depends on tube Rp and geometry. Other tubes will cluster at a different %UL. One will notice that the distortion increases for the lower impedance xfmrs with higher power output. One might want to scale the power curves for constant distortion at max power, then the %UL's for this tube will move around.
If max cathode current is kept the same (another option), then these curves will move toward the triode %UL as the xfmr Zload is lowered in order for the Rp to match the Zload.
Next, I have not advocated lower %UL, perhaps the 20% taps on my Edcor xfmr are confusing the issue, they were for CFB experiments.
If anything I would advocate higher %UL taps in the interest of lower distortion. But max power will likely come down as triode mode is approached. This is a tradeoff for the designer to decide.
On the screen current distortion issue, several factors likely have prevented noticeable distortion problems. First, the distortion is likely to be even harmonic so is eliminated by P-P.
Secondly all the papers quoted so far are using class A operation, which will eliminate most of the generation of odd harmonics by full overlap of the complementary monotonic screen distortion effects. My concern as stated earlier, is that most comercial UL amps do not run in class A, so likely will have increased odd harmonic distortion. Also, there are some SE xfmrs around that have UL taps, I would expect the screen current distortion there to be quite noticeable.
Thirdly, the traditional audio tubes used have HV screen grids with lower transconductance, these generally have much more forgiving screen current curves. The later designed, low voltage screen pentodes are much less forgiving. Many DIY designs use these low cost (well they used to be!) horizontal and vertical TV scan tubes.
And, well, we just want to test this screen current issue to see if it is a problem nowadays, with the greater emphasis placed on avoiding odd harmonics.
And, I wouldn't be surprised if someone has come up with the same effective triode idea before, I just haven't seen anything published yet. Its not really rocket science, I really thought it was obvious, it really is just ordinary feedback theory. Thats why I cannot see H-K not picking up on it in their paper.
Don
Firstly, the full effective triode formulas do not go to infinity at %UL = 0, I just said that the plate term in the formula was small so it could be ignored for most cases and the formulas simplified to 1/UL% form. With the plate term included, the Rp does indeed converge on the pentode Rp and the Mu effective does converge on the pentode effective Mu. Here are the actual formulas:
Mu_eff = -dVp/dVg = 1/ [ (U%/Mu_scrn) + (1/Mu_plate) ]
Rp_eff = dVp/dI = [K*Ik^(-1/3)] / [ (U%/Mu_scrn) + (1/Mu_plate) ]
or: Rp_eff = K * Mu_eff *Ik^(-1/3)
and Gm = Ik^(1/3) / K ( K is a constant for units and tube geometry, Ik is cathode current )
which nicely gives: Mu_eff = Gm * Rp_eff ( just like any triode! )
OK, next on the KT88 curves. The tube Rp is affected by the plate (or cathode) current (see formula above, Rp proportional to Ik^-1/3). These KT88 curves are scaled power-wise so that the same Rp (or Zout on the graph) occurs for each case, hence the inclusion of only one Zout curve on the graph. Due to this, one expects all the curves to peak at the same %UL. This does not tell us much. The info we want has been factored out.
Now it could be that the power curves shown are the maximum power that can be achieved at some max tube dissipation, and then we could conclude that Ik^1/3 conspires to keep the peak at nearly the same %UL for all cases of xfmr Zload for this tube.
But this favored %UL will still be specific to the particular tube chosen because it depends on tube Rp and geometry. Other tubes will cluster at a different %UL. One will notice that the distortion increases for the lower impedance xfmrs with higher power output. One might want to scale the power curves for constant distortion at max power, then the %UL's for this tube will move around.
If max cathode current is kept the same (another option), then these curves will move toward the triode %UL as the xfmr Zload is lowered in order for the Rp to match the Zload.
Next, I have not advocated lower %UL, perhaps the 20% taps on my Edcor xfmr are confusing the issue, they were for CFB experiments.
If anything I would advocate higher %UL taps in the interest of lower distortion. But max power will likely come down as triode mode is approached. This is a tradeoff for the designer to decide.
On the screen current distortion issue, several factors likely have prevented noticeable distortion problems. First, the distortion is likely to be even harmonic so is eliminated by P-P.
Secondly all the papers quoted so far are using class A operation, which will eliminate most of the generation of odd harmonics by full overlap of the complementary monotonic screen distortion effects. My concern as stated earlier, is that most comercial UL amps do not run in class A, so likely will have increased odd harmonic distortion. Also, there are some SE xfmrs around that have UL taps, I would expect the screen current distortion there to be quite noticeable.
Thirdly, the traditional audio tubes used have HV screen grids with lower transconductance, these generally have much more forgiving screen current curves. The later designed, low voltage screen pentodes are much less forgiving. Many DIY designs use these low cost (well they used to be!) horizontal and vertical TV scan tubes.
And, well, we just want to test this screen current issue to see if it is a problem nowadays, with the greater emphasis placed on avoiding odd harmonics.
And, I wouldn't be surprised if someone has come up with the same effective triode idea before, I just haven't seen anything published yet. Its not really rocket science, I really thought it was obvious, it really is just ordinary feedback theory. Thats why I cannot see H-K not picking up on it in their paper.
Don
ummm, Smokingamp...your quote:
Secondly all the papers quoted so far are using class A operation, which will eliminate most of the generation of odd harmonics by full overlap of the complementary monotonic screen distortion effects.
This overlap in Class A PP does in the even. The odd stuff is symetrical( v. the asymetrical 2HD ), is not effected by that operating method.
Cheers,
Douglas
Secondly all the papers quoted so far are using class A operation, which will eliminate most of the generation of odd harmonics by full overlap of the complementary monotonic screen distortion effects.
This overlap in Class A PP does in the even. The odd stuff is symetrical( v. the asymetrical 2HD ), is not effected by that operating method.
Cheers,
Douglas
"This overlap in Class A PP does in the even."
Umm, I said that in the line above what you quoted:
"First, the distortion is likely to be even harmonic so is eliminated by P-P."
What I meant by:
" class A operation, which will eliminate most of the generation of odd harmonics by full overlap of the complementary monotonic screen distortion effects. "
was a second order effect in P-P, where placing even order distortions back to back produces some odd order distortion. The more it is overlapped, as in class A versus class B, the less symmetrical or odd order residual remains.
Basically, averaging the gain of two devices smoothes out the variation. For class B, there is no overlap so no averaging.
Ie, two J shaped curves when placed back to back make an S shaped curve. The more we overlap the J's, the less of an S remains. Overlapping doesn't get rid of all the odd residuals however, particularly not the high order ones. As in most distortion cancellation schemes, the low order cancellations are effective, the high order ones are not.
It's like the LTP or differential stage, the even order distortions cancel, but an S shaped odd order one remains. It just saturates on BOTH ends now.
Don
Umm, I said that in the line above what you quoted:
"First, the distortion is likely to be even harmonic so is eliminated by P-P."
What I meant by:
" class A operation, which will eliminate most of the generation of odd harmonics by full overlap of the complementary monotonic screen distortion effects. "
was a second order effect in P-P, where placing even order distortions back to back produces some odd order distortion. The more it is overlapped, as in class A versus class B, the less symmetrical or odd order residual remains.
Basically, averaging the gain of two devices smoothes out the variation. For class B, there is no overlap so no averaging.
Ie, two J shaped curves when placed back to back make an S shaped curve. The more we overlap the J's, the less of an S remains. Overlapping doesn't get rid of all the odd residuals however, particularly not the high order ones. As in most distortion cancellation schemes, the low order cancellations are effective, the high order ones are not.
It's like the LTP or differential stage, the even order distortions cancel, but an S shaped odd order one remains. It just saturates on BOTH ends now.
Don
6973
Ummm, here's what I came up with:
gm1 = 4800 @ 46mA
gm2 = 30mA/50V = 600
Rscrn = 1/gm2 = 1666
Mu = gm1/gm2 = 8
Hmmm, looks like its a 9 pin version of a 6L6!
6L6: gm1 = 4700
Mu = 8
gm2 = Mu/gm1 = 1700
Rscrn = 1/gm2 = 587.5
Been there, done that!
Well, might be interesting to see if the 6973 UL data matches the 6L6 UL data.
Any UL stuff available on one of the horizontal or vertical output tubes?
Don
Ummm, here's what I came up with:
gm1 = 4800 @ 46mA
gm2 = 30mA/50V = 600
Rscrn = 1/gm2 = 1666
Mu = gm1/gm2 = 8
Hmmm, looks like its a 9 pin version of a 6L6!
6L6: gm1 = 4700
Mu = 8
gm2 = Mu/gm1 = 1700
Rscrn = 1/gm2 = 587.5
Been there, done that!
Well, might be interesting to see if the 6973 UL data matches the 6L6 UL data.
Any UL stuff available on one of the horizontal or vertical output tubes?
Don
smoking-amp said:
Not unless someone traced the curves. It should be possible to calculate the relevant data for sweep tubes that have published triode data, EL504/PL504 would be one like that, but of course, there would be nothing to check them against
As an aside, given that most sweep tubes have (often vastly) reduced maximum Vg2 WRT maximum Vp, it is no surprise 'classical' UL data is not provided, as it is unlikely that the tube can be used with a tapped transformer, unless the power supply is lower than the usual standard values - and we all know that separate G2 windings are not at all common. I think your approach with a MOSFET follower would be of great benefit here.
smoking-amp said:And... we would not be stuck with all these 43% tapped output transformers on the market that are totally meaningless!
Don
To summarise the above: 43% tapped output transformers are
totally meaningless.
Care to explain, in the light of practical measurements (to try to get some practice in here edgewise as you do not seem to have constructed anything so-far - one does want to make one's amplifier produce music, doesn't one?). And you have mentioned several times that the manufacturer's graphs do not give everything - that is not an explanation - they never do! Point is: do they give useful designer's data, or are we busy here with a mainly academic exercise?
But could one for once get to the nitty-gritty before continuing an otherwise very informative subject: If I use manufacturer's data as a first approximation in circuit design, and I find that a spectrum analyser (which is not out of order) gives roughly corresponding data ....
AND... being a researcher, I find that changes from that do not significantly improve my design .......
how must I arrive at the conclusion that "43% taps are meaningless"?? (I think it is the 3rd time I have asked this question in some way or another. I am not trying to be disrespectful in any way, Don, but allow me to also say: Been there, done that! (Didn't get the T-shirt; in those days there were none). Still doing it (still no T-shirts).
I am also not disputing the value of what you are presenting here (I understand that; just for the record, did pass Maths IV at varsity). But either the manufacturer's graphs are misleading or they are useful. If the former, see previous paragraph. If the latter, why do I need mathematical gymnastics in order to design a successful amplifier (and why do you, to my perception, dismiss everything that doesn't agree with you)?
(More on graphs later)
Kind regards.
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