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Class D Switching Power Amplifiers and Power D/A conversion

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Old 20th June 2011, 04:34 PM   #21
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I'm afraid, we DIYers are not the most attractive customers.
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Old 5th December 2011, 10:44 AM   #22
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@ Choco: max dv/dt = 10V/ns from the newest online datasheet
diode Qrr behavior: I am already in contact with one of the sales managers to get more info..
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Old 6th December 2011, 11:16 AM   #23
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Hi Tom,
glad to see that they improve their data sheets.
If 10V/ns are sufficient or not, may depend on the specific application.
In my current design it would not be sufficient.

First I was checking if the IRFB4615 would work for me.
It is a pretty nice and fast switch, but with limited power handling capabilities.
Found that its limitations are in the range of 25-30A, when hard switching with 400kHz.
So I am right now planning to use the IRFB4115, of course they need a stronger gate drive compared vs 4615...
Attached two screen shots when operating the IRFB4115 at 41A continuous current load under hard switching conditions with 500kHz (I usually simulate heavy bass load by loading the half bridge with constant current. Simply a series of some hundrets mOhms + 20uH from half bridge output to positive rail and duty cycle adjusted to get the desired current).
The first step in Uds is caused by the massive di/dt during commutation from upper Mos to lower Mos. This di/dt is above 1000 A/us and causes an inductive drop around 30V. Please note the RdsON and the dead time are chosen in a way that the body diodes do not store much charge. So I do not need to remove full Qrr.
After 41,6ns the current has commutated to the lower switch and the half bridge output jumps within 4ns to the lower rail. The resulting dv/dt is around 15V/ns.


Upper Trace: Ugs of lower Mos, 5V/div
Lower Trace: Uds of lower Mos, 50V/div
Attached Images
File Type: jpg 41A500k.JPG (112.2 KB, 551 views)
File Type: jpg dvdt.JPG (98.0 KB, 526 views)
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Old 6th December 2011, 12:53 PM   #24
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@choco: from my understanding It looks the 41.6ns period is the reverse recovery of the upper diode. The short 4ns jump is the diode's "snap-off" causing high dv / dt. Maybe a snubber can help.
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Old 6th December 2011, 01:56 PM   #25
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The 41.6ns is the time for both.
Taking over the load current and removing Qrr.
First the current in the lower switch rampes up to full load current,
then continuos to ramp up further to remove the Qrr.
Already with Qrr=0, we would need a di/dt around 1000A/us to commutate 40A within 40ns.
If you increase the dead time, then the current will commutate to the body diode right after turning OFF the upper switch but before the lower switch turns ON.
If you avoid long dead times, then the process of becoming non conductive of the upper switch and becoming conductive of the lower switch happen in parallel. The diodes will see just very few charge. The amount of charge which we have to remove is not larger than the amount of charge that was applied during forward time of the diodes.
You need high switching precision for such an adjustment, but it is possible.

If you introduce longer dead time in the above set up, then the time for commutation will increase significantly and the losses too.
Drawback of the chosen dead time setting is that the adjustment, which is perfect at high load, will cause a significant cross conduction peak during idle condition. Overall it is a trade off, which everybody has to settle in his design, according his requirements.
If you optimize for low idle losses, your switches will struggle more with Qrr at high loads....

High dv/dt will happen in any case, the more Qrr you have to remove the more tough. ..oh yes the diode snap off right after Qrr is removed is no fun.

Snubbers could theoretically help to reduce dv/dt even further.
But consider which values you will need.
Let's assume the body diode might have not been fully flooded.
May be the Qrr removing current was just 20A (which are on top of the 40A), then the lower Mosfet will be in the situation of drawing 60A in the moment of diode snap off. 40A will be delivered by the load, the other 20A will have to be catched by the snubbers and parasitic capacitances.
The shown screen shot was taken with 470pF+12 Ohms directly from drain to source at each switch.
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Old 6th December 2011, 08:30 PM   #26
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I forgot to highlight one other effect.
The massive dip in Ugs in the moment of sloping.
Most people think it would be caused by the Miller capacitance.
In fact that's just one portion of the dip.
The larger portion is caused by the huge 'snap off di/dt' of the diode.
The upper body diode snaps off and suddenly the lower Mos does not need to carry the removal current anymore. Means the current in the switch has to come down from i.e. 60A to 40A very fast. Lead inductance of the TO-220 source lead will show a significant voltage drop, which we see as a dip when measuring the Ugs.
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Old 21st December 2011, 02:28 AM   #27
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Excellent work with di/dt!!
Waveforms look quite good. Turn off would be even better with slightly lower gate drive inductance. The glitches in gate drive (due to buffers?) could be smoothed out too.

470pf + 12 ohms seems a bit "light", and ringing looks slightly underdamped. What is the frequency of the major resonant mode that appears without snubber? I get over 100Mhz with IRFB4227 (lower Cds). Did you try to tune C and R parasitic inductance at the main mode and then reduce R until best damping is achieved? A good start value is a C similar or slightly higher than Cds at max. typ turned-off Vds.

Guess what? I have just confirmed that my big sub amps with "crazy switching" pass radiated EN55022B, two dominant peaks appear in the spectrum. One is associated with turn on slopes and the other associated with turn off slopes (its harmonics are slightly noticeable too). A smaller and lower Q peak appears around FET resonant mode too. Small ferrite bead was needed in speaker wires in the end (ceramic chips to case/gnd not practical in this design).
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Last edited by Eva; 21st December 2011 at 02:35 AM.
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Old 21st December 2011, 05:25 PM   #28
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Hi Eva,
my congrats that you passed radiated EMI

Dominant resonant mode without snubbers in this bread board up was below 100MHz. Would need to measure again, but I think it was around 80MHz.
Based on your older hints and findings of my older proto (4 years ago), I also experimented with the geometry of the snubbers, but strangely - this time it showed just a minor effect in this bread board.

470pF are light? Hm, 2 years ago I tweaked my old proto to 1MHz switching frequency and reduced the snubber caps for the IRFB4321, because of heat. There I found a cool snubber geometrie which was perfect with 220pF and approx 10 Ohms. Well, cool from damping effect, but the necessary snubber loop was in the category of 10cm² ! ...having some doubts that this would be fortunate for radiated EMI.

Lower gate resistor for turning OFF:
It seems that I am already touching the limitations of the parasitic inductance of my gate drive loop in that proto.

Glitches:
The glitches at the end of turning OFF seem to be related to the reverse recovery of the boot strap supply diode. Originally it was much larger. Changing the charge pump wiring and also optimizing the boot strap diode reduced the glitches to a reasonable level.

...let's see how things will come with a proper PCB.
This time I am definitely on the trip to settle a nice amp with double layer PCB, in order to play around with a new gain and loop topology that overcomes a good portion of the distortions, which are related to the carrier residuals in clocked post filter designs. In simulation it appears great, time to go for the real thing. Playing with KiCad these days...

Tom:
Sorry for 'off topicking' your thread.
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Old 21st December 2011, 10:53 PM   #29
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Concerning turn off, I meant reducing loop inductance, not the resistance.

With "light" I meant too high resistance, capacitance is probably ok.
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Old 22nd December 2011, 12:35 AM   #30
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Quote:
Originally Posted by Eva View Post
Diode forward and reverse recovery processes are symmetrical. The charge that takes 100ns to be removed from the diode took another 100ns to be stored... Oh, wait, was the diode allowed to conduct full output current for so long? Obviously no.
Obviously not.

In fact, it can be worse than that. Far worse.

When a diode is forward-biased at high dI/dt, it can develop a forward voltage of 20, 40, even 100 volts at very high rates -- and this is a real voltage applied to the junction, it is not due to lead inductance, which does increase the apparent measurement, but does not account for the whole thing. Quite simply, the instant you apply voltage, there's no charge in the diode (assuming it's initially off), so the voltage can be pretty much anything until charge carriers flood across the junction and it becomes conductive.

When forward-biased suddenly, charge drifts into the junction, and it gradually becomes more conductive. If you reverse-bias it within the transit time, you can generate clouds of charge within the junction. Even if you simply reverse-bias a quiescent (fully saturated, steady state, whatever) junction, you can get behavior where regions of the junction deplete sooner than others.

When packets of charge occur, the junction is effectively shorted out in part. If the other part comes out of recovery, it becomes an insulator. Now the full reverse voltage appears across a narrow thickness of the junction, and the avalanche voltage is effectively reduced -- this can lead to a phenomenon known as dynamic avalanche, where it might break down (causing excessive losses and ringing) at a fraction of the rated voltage. This is particularly important under conditions of high voltage, high dI/dt and large stray inductance, which are precisely the conditions you'll get under arbitrary load current, short deadtime conditions.

Now, power diodes don't usually have the same doping profiles as SRDs (step-recovery diodes), but under the right conditions they can behave in exactly the same way. This can lead to incredibly high dV/dt recovery slopes, resulting in excessive EMI and potentially dangerous peak voltages. The trigger is usually short on-times and high dI/dt recovery conditions.

The biggest issue is high voltage diodes, where the junction width is necessarily large. This is a concern to myself, because I regularly work with 480VAC input circuitry, which typically requires 1200V rated diodes, which have wide junctions, giving ample time for transition time effects to take effect.

On the plus side, 200V diodes are substantially faster (and there are silicon schottkies available, completely obviating this operating condition), which makes class D operation easier. Still, it won't completely go away; forward and reverse peak voltages are always a concern, and small changes can cause large EMI problems. The best solution is to always be aware of these limits and design accordingly. You can usually do better by reducing stray inductance and capacitance, but increased stray inductance and snubber capacitance often improve switching, rather than impairing it as many believe.

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