Aletheian,
You have had so much good advice that you hardly need more, but since I seem to be writing nothing tonight....
I would echo PRR (post #15) and just use the circuit without NFB for your purpose. Otherwise a no go; you will need an input stage for anything proper. I have seen many attempts over a long time at making an over-simple circuit too prosperous - often not worth the effort because of minimal real advantage, if at all. As was said, straight output triodes are .... mmm ... not too bad on their virgin own.
You have had so much good advice that you hardly need more, but since I seem to be writing nothing tonight....
I would echo PRR (post #15) and just use the circuit without NFB for your purpose. Otherwise a no go; you will need an input stage for anything proper. I have seen many attempts over a long time at making an over-simple circuit too prosperous - often not worth the effort because of minimal real advantage, if at all. As was said, straight output triodes are .... mmm ... not too bad on their virgin own.
While on the subject of Cathodyne Phase Splitters can anyone tell me what the effective source resistance (impedance) is in terms of tube parameters and RL = RK resistance value.
The 1960 article by Albert Priesman seems to suggest that the way to think of this is that the "effective" source resistance at both anode and cathode is rp + (1+mu)(RL+RK) so long as RL=RK and the effective source voltage is considered to be mu x Vgk.
Since we know that the gain of this thing is actually close to 1 which implies an effective source voltage approx. equal to vgk this requires some mental gymnastics to visualise.
Considering it as an approximately unity gain circuit and adjusting the above to suit would suggest that the effective source impedance is approx rp/mu + RL + RK OR 1/gm + RL + RK
Can anyone confirm the "maths interpretation" of what Priesman wrote and/or my interpretation of the same.
I have a Cathodyne driving 2 pairs of EL34 in Triode Mode. To improve the drive (at high frequencies in particular) I need a lower effective source impedance.
That would suggest either ECC99 (what I'm using now) or 6H30 at higher current setting and correspondingly reduced RL and RK is the way to go. This would seem to be right intuitively.
Does this seem right to you experts or have I missunderstood the Preisman article or screwed up the maths.
Should I perhaps just forget the B maths and analysis - stick on the bench, make the changes and see what happens?
Cheers,
Ian
The 1960 article by Albert Priesman seems to suggest that the way to think of this is that the "effective" source resistance at both anode and cathode is rp + (1+mu)(RL+RK) so long as RL=RK and the effective source voltage is considered to be mu x Vgk.
Since we know that the gain of this thing is actually close to 1 which implies an effective source voltage approx. equal to vgk this requires some mental gymnastics to visualise.
Considering it as an approximately unity gain circuit and adjusting the above to suit would suggest that the effective source impedance is approx rp/mu + RL + RK OR 1/gm + RL + RK
Can anyone confirm the "maths interpretation" of what Priesman wrote and/or my interpretation of the same.
I have a Cathodyne driving 2 pairs of EL34 in Triode Mode. To improve the drive (at high frequencies in particular) I need a lower effective source impedance.
That would suggest either ECC99 (what I'm using now) or 6H30 at higher current setting and correspondingly reduced RL and RK is the way to go. This would seem to be right intuitively.
Does this seem right to you experts or have I missunderstood the Preisman article or screwed up the maths.
Should I perhaps just forget the B maths and analysis - stick on the bench, make the changes and see what happens?
Cheers,
Ian
Hi,
I have both measured and simulated the Cathodyne Phase Splitter and have confirmed that the Preisman calculations are correct, it is not that difficult to do.
However given that the gain to each output is normally given as u/(u+1) the output impedance will rather be (rp+Zk)/(u+1) i.e. much lower than you would expect, (it is described in the article, look at figure 3 and the surrounding text).
Do what I did, do some simulations in Spice or build one or 2 phase splitters and measure and you will find the results I did.
The reason I did this was because I earlier measured the open loop response of my OTL and didn't discover the poles I expected, the poles due to the phase splitter was much higher which of course was due to that the output impedance of the phase splitter was much lower than what I assumed earlier.
My measurementy method was to use different values of load impedances, (but equal for cathode and anode outputs) and comparing the voltage, from the voltage difference I calculated the output impedance.
The confusion around the cathodyne is probabaly due to that some want to measure the output impedance of cathode and anode independantly and then they are indeed very different, however when the load impedances are seen as independant it is wrong to talk about this circuit as a phase splitter, isn't it?
To your questions:
You conclusions are correct but only if you not consider that that particular definition of output impedance also implies that the gain is u*Z/(rp+(1+u)Z+Z).
Given that the gain is most often written as u/(u+1) and the output impdance as I wrote above as (rp+Zk)/(u+1) you can draw the following conclusions:
rp is as critical as the anode or cathode resistors
High u is beneficial, (as long as the load impdances are equal but lower u keeps better balance if the loads are unbalanced, e.g class A2)
I use 12BH7 or 6SN7 as phase splitter as I think that fulfill the above requirements to a reasonable compromise.
Regards Hans
I have both measured and simulated the Cathodyne Phase Splitter and have confirmed that the Preisman calculations are correct, it is not that difficult to do.
However given that the gain to each output is normally given as u/(u+1) the output impedance will rather be (rp+Zk)/(u+1) i.e. much lower than you would expect, (it is described in the article, look at figure 3 and the surrounding text).
Do what I did, do some simulations in Spice or build one or 2 phase splitters and measure and you will find the results I did.
The reason I did this was because I earlier measured the open loop response of my OTL and didn't discover the poles I expected, the poles due to the phase splitter was much higher which of course was due to that the output impedance of the phase splitter was much lower than what I assumed earlier.
My measurementy method was to use different values of load impedances, (but equal for cathode and anode outputs) and comparing the voltage, from the voltage difference I calculated the output impedance.
The confusion around the cathodyne is probabaly due to that some want to measure the output impedance of cathode and anode independantly and then they are indeed very different, however when the load impedances are seen as independant it is wrong to talk about this circuit as a phase splitter, isn't it?
To your questions:
You conclusions are correct but only if you not consider that that particular definition of output impedance also implies that the gain is u*Z/(rp+(1+u)Z+Z).
Given that the gain is most often written as u/(u+1) and the output impdance as I wrote above as (rp+Zk)/(u+1) you can draw the following conclusions:
rp is as critical as the anode or cathode resistors
High u is beneficial, (as long as the load impdances are equal but lower u keeps better balance if the loads are unbalanced, e.g class A2)
I use 12BH7 or 6SN7 as phase splitter as I think that fulfill the above requirements to a reasonable compromise.
Regards Hans
Hans,
THANK YOU !!!
So I want higher mu, lower RK and to a lesser extent lower rp (since for any sensible tube selection rp will be 1/4 off RK or less)
The higher mu will be handy since the whole amp is a bit short of gain and the other half of the tube is the input common cathode amp.
So I'll try
1) 6N1P (mu=35, rp = 8K)
2) 12AT7 (mu=70, rp =8K)
and drop the anode and cathode load resistors to the extent possible consistent with conservative power dissipation limits. Actually will try the 12AT7 first since that doesn't need any heater rewiring.
Have ONLY last night installed B2 Spice on my home machine so may try simulating as well.
Cheers,
Ian
THANK YOU !!!
So I want higher mu, lower RK and to a lesser extent lower rp (since for any sensible tube selection rp will be 1/4 off RK or less)
The higher mu will be handy since the whole amp is a bit short of gain and the other half of the tube is the input common cathode amp.
So I'll try
1) 6N1P (mu=35, rp = 8K)
2) 12AT7 (mu=70, rp =8K)
and drop the anode and cathode load resistors to the extent possible consistent with conservative power dissipation limits. Actually will try the 12AT7 first since that doesn't need any heater rewiring.
Have ONLY last night installed B2 Spice on my home machine so may try simulating as well.
Cheers,
Ian
After that good analysis by Hans perhaps just a practical point, i.e. it must be kept in mind that rp can change with the load resistors in the sense that it is dependant on Ia. I also like using the ECC81 here, but unfortunately do not have parameter/Ia graphs for that. Perhaps an illustration using ECC88 parameters might be interesting (mu about 29 and Gm about 3 mA/V here):
For a tube Va-k of 150V (where my graphs are valid) and h.t. of 400V, it is found that:
For RL=RK=33K, Ia=4.4 mA and rp=5.5K
For RL=RK=47K, Ia=2.6 mA and rp=8.5K
For RL=RK=100K (rather high), Ia=1.3 mA and rp=12K
This means that the (rp+Rk) term does not change all that much.
But Hans: You conclude that rp is as critical as RK (or RL), but according to the above it forms about 12% - 18% only of RK (RL). Do I misunderstand you here?
Also, Gingertube, the ECC81 has a mu of about 57 and rp of about 18K at the currents normally used, which make things a bit more critical - although, as said, I have often used it (I normally do not drive output triodes directly from a cathodyne, which eases matters somewhat for me).
And what about the E182CC? It can be used at quite a high current (mu about 22 and rp about 3K at 9 mA). One can of course go on suggesting tubes ad infinitum.....
For a tube Va-k of 150V (where my graphs are valid) and h.t. of 400V, it is found that:
For RL=RK=33K, Ia=4.4 mA and rp=5.5K
For RL=RK=47K, Ia=2.6 mA and rp=8.5K
For RL=RK=100K (rather high), Ia=1.3 mA and rp=12K
This means that the (rp+Rk) term does not change all that much.
But Hans: You conclude that rp is as critical as RK (or RL), but according to the above it forms about 12% - 18% only of RK (RL). Do I misunderstand you here?
Also, Gingertube, the ECC81 has a mu of about 57 and rp of about 18K at the currents normally used, which make things a bit more critical - although, as said, I have often used it (I normally do not drive output triodes directly from a cathodyne, which eases matters somewhat for me).
And what about the E182CC? It can be used at quite a high current (mu about 22 and rp about 3K at 9 mA). One can of course go on suggesting tubes ad infinitum.....
Experimental Results from last night
The amp in question has a 12AU7 Common Cathode / Direct coupled Cathodyne feeding 2 pairs of triode strapped EL34s into a Plitron PAT4006 (VDV2100) Toroidal O/P Transformer (1900raa : 5 secondary). Cathodyne RA and RK are 22K and stage current is 4.4mA.
NOTE ALL Measurements except max power refered to 6V RMS across 3.5 Ohm load @ 1kHz (approx 10W out) - 0.8V RMS input. These levels set using my multimeter.
Baseline measurements with 12AU7:
Power into 3.5 Ohm dummy load @ 1KHz = 39W
Zout @10W, 1kHz = 4.4 Ohms
Gain = 5.5
High Frequency -3dB point is 225kHz
Assuming approx 100pF each side for the parallel EL34 inputs that tells me straight away that the effective source impedance from the cathodyne is NOT MORE than 7KOhms.
Change to ECC99 - Sound was a lot cleaner (The Common Cathode stage???)
Power Out - unchanged
Zout - some suggestion it was slightly higher at 1kHz (measured a value of 4.9 - shifted pole from cathodyne effective source Z ???)
Gain = 7.2
High Frequency -3dB point is 250kHz
While running the frequency response checks using oscilloscope I could see the effect of the feedback from the Miller capacitance of the EL34s. As frequency went up, so did feedback due to Miller Cap. This reduced EL34 rp which reflected to the secondary as reduced Zout and we saw a boost in voltage across the 3.5Ohm dummy load.
Zout measures (Vout open circuit =88V p-p):
Freq. Vout Zout
20 28 7.5
50 34 5.6
100 35 5.3
200 35 5.3
500 35.5 5.2
1000 37 4.8
2000 40 4.2
5000 46 3.2
10000 48 2.9
20000 49 2.9
50000 49 2.9
100000 42 3.8
200000 28 7.5
These were a consistent set of readings from the oscilloscope so Zout results are correct BUT for some reason it doesn't stack up with the 6V RMS into 3.5 Ohms @ 1KHz set using my multimeter. 6V RMS is NOT 37 V pk-pk????
I havent bothered BUT I think that you could calculate backward from this data to arrive at an effective Source impedance from the cathodyne.
Also: There was some indication (on the oscilloscope) of slewrate limiting starting at about 50kHz
The gain difference 7.2 vs 5.5 can be explained by difference in mu (22 vs 17).
I think the next step for me and this amp is ECC81 front end with 6dB of global feedback.
Hope there is stuff of interest to you in the data.
Cheers,
Ian
The amp in question has a 12AU7 Common Cathode / Direct coupled Cathodyne feeding 2 pairs of triode strapped EL34s into a Plitron PAT4006 (VDV2100) Toroidal O/P Transformer (1900raa : 5 secondary). Cathodyne RA and RK are 22K and stage current is 4.4mA.
NOTE ALL Measurements except max power refered to 6V RMS across 3.5 Ohm load @ 1kHz (approx 10W out) - 0.8V RMS input. These levels set using my multimeter.
Baseline measurements with 12AU7:
Power into 3.5 Ohm dummy load @ 1KHz = 39W
Zout @10W, 1kHz = 4.4 Ohms
Gain = 5.5
High Frequency -3dB point is 225kHz
Assuming approx 100pF each side for the parallel EL34 inputs that tells me straight away that the effective source impedance from the cathodyne is NOT MORE than 7KOhms.
Change to ECC99 - Sound was a lot cleaner (The Common Cathode stage???)
Power Out - unchanged
Zout - some suggestion it was slightly higher at 1kHz (measured a value of 4.9 - shifted pole from cathodyne effective source Z ???)
Gain = 7.2
High Frequency -3dB point is 250kHz
While running the frequency response checks using oscilloscope I could see the effect of the feedback from the Miller capacitance of the EL34s. As frequency went up, so did feedback due to Miller Cap. This reduced EL34 rp which reflected to the secondary as reduced Zout and we saw a boost in voltage across the 3.5Ohm dummy load.
Zout measures (Vout open circuit =88V p-p):
Freq. Vout Zout
20 28 7.5
50 34 5.6
100 35 5.3
200 35 5.3
500 35.5 5.2
1000 37 4.8
2000 40 4.2
5000 46 3.2
10000 48 2.9
20000 49 2.9
50000 49 2.9
100000 42 3.8
200000 28 7.5
These were a consistent set of readings from the oscilloscope so Zout results are correct BUT for some reason it doesn't stack up with the 6V RMS into 3.5 Ohms @ 1KHz set using my multimeter. 6V RMS is NOT 37 V pk-pk????
I havent bothered BUT I think that you could calculate backward from this data to arrive at an effective Source impedance from the cathodyne.
Also: There was some indication (on the oscilloscope) of slewrate limiting starting at about 50kHz
The gain difference 7.2 vs 5.5 can be explained by difference in mu (22 vs 17).
I think the next step for me and this amp is ECC81 front end with 6dB of global feedback.
Hope there is stuff of interest to you in the data.
Cheers,
Ian
Hi,
Johan, I agree with your post, as rp is dependant on the operating point you have to be a bit careful when selecting anode/cathode resistors.
No it is not a misunderstanding but rather me being incomplete or unclear with my statement, it depends on what you mean by either rp or RK being most critical, what I meant is that a reduction of rp in ohms is as beneficial as the same reduction of RK, of course you could say that it is better to compare for the same reduction in % and in that case it would be more beneficial to reduce RK rather than rp, RK being larger.
Gingertube, I think you have some interesting results, a 12AU7 at 4.4mA should have a rp of about 13kohm and a mu of about 16 which should give a Zout of ~(13 + 22)/(16+1) or ~2.1kohm which woul give 2 poles close to 760kHz assuming 100pF in input capacitance of the EL34's. Changing to ECC99 but keeping the 22k's would then not change the poles so much but maybe a ECC99 allow for lower RK's as well?
BTW, 6VRMS would be 2*SQR2*6 or ~16.9V Pk-Pk
Regards Hans
Johan, I agree with your post, as rp is dependant on the operating point you have to be a bit careful when selecting anode/cathode resistors.
No it is not a misunderstanding but rather me being incomplete or unclear with my statement, it depends on what you mean by either rp or RK being most critical, what I meant is that a reduction of rp in ohms is as beneficial as the same reduction of RK, of course you could say that it is better to compare for the same reduction in % and in that case it would be more beneficial to reduce RK rather than rp, RK being larger.
Gingertube, I think you have some interesting results, a 12AU7 at 4.4mA should have a rp of about 13kohm and a mu of about 16 which should give a Zout of ~(13 + 22)/(16+1) or ~2.1kohm which woul give 2 poles close to 760kHz assuming 100pF in input capacitance of the EL34's. Changing to ECC99 but keeping the 22k's would then not change the poles so much but maybe a ECC99 allow for lower RK's as well?
BTW, 6VRMS would be 2*SQR2*6 or ~16.9V Pk-Pk
Regards Hans
Hans,
The one major thin I learned from the experiments last night is that the effective source impedance of the cathodyne splitter is NOT a problem in this amp. It is quite adequately driving the input capacitance of the EL34s. This was worthwhile learning as I'd prviously assumed it would be a problem and I was all set to replace the entire front end.
The major problem with the amp is its Zout - thats why I'm now thinking about some global feedback.
I had thought to manipulate the tube type and RL,RK values to get the Zout of the splitter and the output tubes input capacitance to be my dominant pole and aim to set it at about 100kHz (about 1/2 the output tranny -3dB point). This now seems to be less practical than I'd hoped (since the cathodyne output Z is much lower than I expected) so may have to shift the dominany pole to the Common Cathode Stage. The one good thing about the Plitron output tranny is that its bandwidth is so high that ensuring stability is not a difficult thing.
I'm surprised at how good the amps sound into my nominally 6 Ohm speakers (DF=1.3). There is no evidence of bloated or wooly bass, it sounds a little bright but not objectionablly so.
Cheers,
Ian
The one major thin I learned from the experiments last night is that the effective source impedance of the cathodyne splitter is NOT a problem in this amp. It is quite adequately driving the input capacitance of the EL34s. This was worthwhile learning as I'd prviously assumed it would be a problem and I was all set to replace the entire front end.
The major problem with the amp is its Zout - thats why I'm now thinking about some global feedback.
I had thought to manipulate the tube type and RL,RK values to get the Zout of the splitter and the output tubes input capacitance to be my dominant pole and aim to set it at about 100kHz (about 1/2 the output tranny -3dB point). This now seems to be less practical than I'd hoped (since the cathodyne output Z is much lower than I expected) so may have to shift the dominany pole to the Common Cathode Stage. The one good thing about the Plitron output tranny is that its bandwidth is so high that ensuring stability is not a difficult thing.
I'm surprised at how good the amps sound into my nominally 6 Ohm speakers (DF=1.3). There is no evidence of bloated or wooly bass, it sounds a little bright but not objectionablly so.
Cheers,
Ian
SY,
Good idea - unfortunately the Plitron PAT4006 (VDV2100) has ONLY a single 5 Ohm Output winding with no centre tap etc.
I do have a pair of VDV2100-CFB/H sitting on the shelf as well, which have both a centre tapped 5 Ohm secondary AND 2 separate 20 Ohm cathode feedback windings BUT I am saving those for the next project where I want to mess about with Ultralinear + Cathode feedback (Menno's "Super Triode" connection).
Learning some good stuff from this lot though.
Thanks,
Ian
Good idea - unfortunately the Plitron PAT4006 (VDV2100) has ONLY a single 5 Ohm Output winding with no centre tap etc.
I do have a pair of VDV2100-CFB/H sitting on the shelf as well, which have both a centre tapped 5 Ohm secondary AND 2 separate 20 Ohm cathode feedback windings BUT I am saving those for the next project where I want to mess about with Ultralinear + Cathode feedback (Menno's "Super Triode" connection).
Learning some good stuff from this lot though.
Thanks,
Ian
I worked through Morgan Jones treatment last night. Her is the summary of his work.
For equal loads on Anode and cathode
Zout = RL.ra/RL(u+2)+ra
The ra term on the bottom line is insignificant compared to RL(u+2) term so drop it. Then the RL terms top and bottom lines cancel leaving
Zout approx = ra/u+2
At typical values of u (>=20) U+2 approx = u, so simplify again
Zout approx = ra/u = 1/gm
If driving output stage directly The equal loads on Anode and Cathode will NOT be guaranteed if:
1) Output stage strays out of Class A (When a tube cuts off it has no gain so Miller capictance will change, particularly with Triode Mode Output, less so with Ultralinear and less so again in Pentode Mode)
2) Output Stage strays into grid current.
If the Anode load drops significantly then:
Zout cathode = RL+ra/(u+2) x ra/RL - ra/RL term insignigicant so
Zout cathode approx = RL+ra/u+2 - at uual values of u
Zout cathode approx = RL/u + ra/u = RL/u + 1/gm
That is it increases by RL/u
If the cathode load drops sifgnificantly then:
Zout anode = RLxRL(u+1)+RL.ra / RL(u+2)+ra
RL squared (u+1) is much larger than RL.ra and RL(U+2) is much larger than ra so
Zout anode approx = RLxRL(u+1)/RL(u+2)
and at reasonable values of u
Zout anode approx = RL
Summary:
As the loads on Anode and cathode become unbalanced then
Zout anode increases from 1/gm toward RL
Zout cathode increase from 1/gm by maximum factor of RL/u
Thats why its suggested that low u is better when driving other than Class A Output Stage.
Cheers,
Ian
For equal loads on Anode and cathode
Zout = RL.ra/RL(u+2)+ra
The ra term on the bottom line is insignificant compared to RL(u+2) term so drop it. Then the RL terms top and bottom lines cancel leaving
Zout approx = ra/u+2
At typical values of u (>=20) U+2 approx = u, so simplify again
Zout approx = ra/u = 1/gm
If driving output stage directly The equal loads on Anode and Cathode will NOT be guaranteed if:
1) Output stage strays out of Class A (When a tube cuts off it has no gain so Miller capictance will change, particularly with Triode Mode Output, less so with Ultralinear and less so again in Pentode Mode)
2) Output Stage strays into grid current.
If the Anode load drops significantly then:
Zout cathode = RL+ra/(u+2) x ra/RL - ra/RL term insignigicant so
Zout cathode approx = RL+ra/u+2 - at uual values of u
Zout cathode approx = RL/u + ra/u = RL/u + 1/gm
That is it increases by RL/u
If the cathode load drops sifgnificantly then:
Zout anode = RLxRL(u+1)+RL.ra / RL(u+2)+ra
RL squared (u+1) is much larger than RL.ra and RL(U+2) is much larger than ra so
Zout anode approx = RLxRL(u+1)/RL(u+2)
and at reasonable values of u
Zout anode approx = RL
Summary:
As the loads on Anode and cathode become unbalanced then
Zout anode increases from 1/gm toward RL
Zout cathode increase from 1/gm by maximum factor of RL/u
Thats why its suggested that low u is better when driving other than Class A Output Stage.
Cheers,
Ian
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