Back in the '30s, according to S. Okamura in the book "History of Electron Tubes", several techniques were invented to linearize vacuum tube gain circuits. Then, shortly after, negative feedback theory was invented, and these linearization ideas were forgotten. Long forgotten! One of these ideas is used, in slightly altered form, in modern IC circuitry design extensively today. And most people think of it as having been invented in the '60s. It was never patented however, for good reason. Today, this technology is called the current mirror (not the same as your usual current source by the way). And the current mirror can be configured to give linear current gain as well as unity gain. One never sees this circuit in tube designs today, despite all the talk of linearizing the circuit before resorting to NFB. So, its about time it got some publicity. Attached, and in the following posts are some schematic diagrams and explanation.
Attachments
Attached are .gif images of schematics for various current mirrors including
a vacuum tube design.
CurrentMirror1 -- a bipolar current mirror with current gain of two, higher
current gain with use of more output transistors (or just use a physically
BIGGER trans. for output).
CurrentMirror2 -- bipolar current mirror, improved linearity via emitter
degeneration resistors.
CurrentMirror3 -- The Wilson current mirror, one of many variations on the
basic current mirror to improve some characteristic or other.
VTCurrentMirror -- the basic Vacuum Tube current mirror, the number of
diodes in series will vary depending on output tube geometry and diode
construction geometry (cathode area, and cathode to plate spacing are the
important parameters), R1 and R2 are optional for increased linearity, their
ratio must be in inverse accordance with the basic current gain ratio
produced by the diode/pentode so as to give equal voltage drops in
operation.
VTCurrentMirror2 -- just some biasing applied to the basic design
note: MosFets can be used in current mirrors also and their circuits are
similar in concept to the Vacuum Tube design
The basic VT Current Mirror operation is to use the diodes as a 2/3 power
non-linear resistor for input current to drive voltage conversion,
the pentode then does 3/2 power drive voltage to output current conversion,
with a net result of linear current gain. It looks simple in principle, but
a number of complexities come into play in practice. The current gain is
determined by geometric considerations between the construction of the
diodes and the pentode. Since 2/3 power and 3/2 power are non-linear
devices, they must also operate on corresponding identical (or inverse
actually) sections of their curves in order to correct each other. This
means not only the right bias points but the scaling of the two devices must
match. This scaling is determined geometrically, so not just any diode will
work with any pentode. In practice, this means stacking diodes in series to
get a close match of the geometric scaling for the pentode used. This can
require anywhere from one to a dozen or more diodes in series depending on
the tubes used. Considerable experimentation is mandatory, including a
variable bias supply and curve tracer or X-Y scope to look at current in
versus current out with a swept current input waveform. In addition, the
geometric spacing of diodes I have tried tend to be rather loosely
controlled in production, so just changing tubes with "identical" tubes can
change the number of series diodes required. A practical approach is to use
a few multi-diode tubes like 6JU8 and provide a jumper to select the number
of sections in series. Obviously, this will NEVER catch on for production
circuit designs, unless some special diodes are designed with strict control
of geometry. But for DIY, anything goes! By the way, using a diode
connected triode or pentode in place of the diodes would work in theory, and
even would allow convenient adjustment of the geometry factor by grid bias,
but in practice, the "inselbildung" or grid "island effect" ruins the
required curvature at small currents.
a vacuum tube design.
CurrentMirror1 -- a bipolar current mirror with current gain of two, higher
current gain with use of more output transistors (or just use a physically
BIGGER trans. for output).
CurrentMirror2 -- bipolar current mirror, improved linearity via emitter
degeneration resistors.
CurrentMirror3 -- The Wilson current mirror, one of many variations on the
basic current mirror to improve some characteristic or other.
VTCurrentMirror -- the basic Vacuum Tube current mirror, the number of
diodes in series will vary depending on output tube geometry and diode
construction geometry (cathode area, and cathode to plate spacing are the
important parameters), R1 and R2 are optional for increased linearity, their
ratio must be in inverse accordance with the basic current gain ratio
produced by the diode/pentode so as to give equal voltage drops in
operation.
VTCurrentMirror2 -- just some biasing applied to the basic design
note: MosFets can be used in current mirrors also and their circuits are
similar in concept to the Vacuum Tube design
The basic VT Current Mirror operation is to use the diodes as a 2/3 power
non-linear resistor for input current to drive voltage conversion,
the pentode then does 3/2 power drive voltage to output current conversion,
with a net result of linear current gain. It looks simple in principle, but
a number of complexities come into play in practice. The current gain is
determined by geometric considerations between the construction of the
diodes and the pentode. Since 2/3 power and 3/2 power are non-linear
devices, they must also operate on corresponding identical (or inverse
actually) sections of their curves in order to correct each other. This
means not only the right bias points but the scaling of the two devices must
match. This scaling is determined geometrically, so not just any diode will
work with any pentode. In practice, this means stacking diodes in series to
get a close match of the geometric scaling for the pentode used. This can
require anywhere from one to a dozen or more diodes in series depending on
the tubes used. Considerable experimentation is mandatory, including a
variable bias supply and curve tracer or X-Y scope to look at current in
versus current out with a swept current input waveform. In addition, the
geometric spacing of diodes I have tried tend to be rather loosely
controlled in production, so just changing tubes with "identical" tubes can
change the number of series diodes required. A practical approach is to use
a few multi-diode tubes like 6JU8 and provide a jumper to select the number
of sections in series. Obviously, this will NEVER catch on for production
circuit designs, unless some special diodes are designed with strict control
of geometry. But for DIY, anything goes! By the way, using a diode
connected triode or pentode in place of the diodes would work in theory, and
even would allow convenient adjustment of the geometry factor by grid bias,
but in practice, the "inselbildung" or grid "island effect" ruins the
required curvature at small currents.
Attachments
I use an assortment of fixtures with a bunch of diode tubes wired in series with selectable series tap points. 6JU8 and 9006 are suitable diode tubes for making current mirrors with gain. A high gm pentode works best for high current gain. Its also possible to use the diode corrector idea on triodes with fixed impedance loads. (triodes are only strictly linear in theory with an infinite impedance load, the corrector allows linearity with a realistic load) Just requires additional resistance in the diode string to compensate for the plate voltage feedback in the triode. Generally, the triode correctors require a LOT of diodes in series however.
Attachments
Just so that you don't think nobody's listening...
A fascinating series of posts. As you say, current mirrors are very popular in the input differential pair of IC op-amps, and with good reason. However, their use also gives a clue to their main problem. IC op-amps rely on wrapping plenty of negative feedback around a high gain amplifier. In other words, the signal levels at the input stage are very small - and that's just as well because the semiconductor current mirrors restrict voltage swing to a few hundred millivolts peak to peak. (I experimented with a semiconductor current mirror on top of an ECC83 and was painfully reminded of the fact.)
I'm interested by your thermionic current mirror with diodes and pentode, as with a suitable choice of pentode and operating point, it should be possible to achieve quite a reasonable voltage swing. Have you made any measurements of swing?
A fascinating series of posts. As you say, current mirrors are very popular in the input differential pair of IC op-amps, and with good reason. However, their use also gives a clue to their main problem. IC op-amps rely on wrapping plenty of negative feedback around a high gain amplifier. In other words, the signal levels at the input stage are very small - and that's just as well because the semiconductor current mirrors restrict voltage swing to a few hundred millivolts peak to peak. (I experimented with a semiconductor current mirror on top of an ECC83 and was painfully reminded of the fact.)
I'm interested by your thermionic current mirror with diodes and pentode, as with a suitable choice of pentode and operating point, it should be possible to achieve quite a reasonable voltage swing. Have you made any measurements of swing?
operational info
"Have you made any measurements of swing?"
Yes, first off, one must get into the mindset of using current levels when using this circuit rather than voltage. Its not a suitable circuit for a high impedance voltage input stage, but more suited to intermediate or even output stages of an amplifier. So, first off, the input of an amplifier would be a voltage to current stage, at small input signal levels, before entering any current mirror stages.
The non-linear resistance on the input is not nearly as bad as the exponential non-linearity in SS current mirrors, so actual scope measurements on the input will show some voltage swing, in fact, this swing will obviously match the operating voltage swing of the pentode grid in a given design. The SS current mirrors are much more tricky to get accurate matching because of the exponential characteristic, so usually only appear in ICs where identical transistors are readily available. The V tube current mirror is actually more practical for discrete component design due to its more gradual 3/2 power law. (and a trimpot in the circuit to set grid bias level and a selectable number of diodes in series and a fair amount of patience on the designer's part!)
Since a constant current gain is enforced, the input current will be simply related to the output current by this gain factor.
I have gotten current gains up to around 97 using a 9KC6 pentode (+13V on G3, 100V on G2) and 6x series 6JU8 diode sections. Using 9006 diodes about doubles the gain yet versus the 6JU8 diodes. What voltage appears on the output of course depends on the load resistance (at least for a pentode with sufficiently high output resistance spec) times the current.
Higher current gain comes from using a smaller diode and higher gm pentode. Too small a diode however may get one into noise problems if the current draw is too small. The more series diodes one has to use to get the scaling correct, the higher the input (avg.) impedance will be and hence the input voltage swing for a given input current swing, but this is not a freely adjustable parameter.
When thinking in current gain mode, instead of voltage gain mode, one actually would prefer the input node to be low impedance so it doesn't swing around and affect the previous stage's output resistance. Only the last output stage reverts back to voltage swing into a load. So an "ideal" current mirror amplifier would see no voltage signals whatsoever internally (Notice that this KILLS Miller effects nicely!) , just on the output. Of course, having a pentode for output with very high output impedance is another problem, so usually one would put a cathode follower, or similar, on the output to lower output impedance or use negative feedback to lower Zout.
"diode(s)) to correct for the similarly non-linear curve of a tube"
Exactly. The diode is a 2/3 power law non-linear resistor and the pentode is a 3/2 power law non-linear Gm. The output is the 3/2 power function of the 2/3 power function of the input. For a function of a function, the exponents get multiplied (rather than added as for a multiplication function). More intuitively, for a given pentode grid voltage, the diode's input current will be the 3/2 power of the grid voltage and likewise the pentode's output current will be the 3/2 power of the grid voltage. So input to output current ratio is linear. The current gain comes simply from geometric scaling of the devices, use a bigger pentode (or higher gm) and you get more current out for the same grid voltage. Use a smaller diode and you get less input current.
This current mirror operation is incidentally quite similar to positive grid operation of a triode or pentode, everything just built into the same tube then. Only problem is the diode (grid) current subtracts from the plate current, eventually hogging the cathode current and causing saturation. So, not as linear as the current mirror configuration.
Another variation on this current mirror idea, (in fact it was invented this way I think) is to reverse the order. Pentode first, then a big diode for load impedance. This is the voltage mirror, and gives linear voltage gain. Unfortunately, its not too practical, as the load is the diode, nothing else allowed except maybe just a grid load. Not to mention that it takes a big diode to correct a small pentode. (clunk -- tossing into the dustbin)
Another useful application of the current mirror could be as a linearity corrector for a SET amplifier, the diode input corrector (with additional resistor) gets applied to the input of the triode. This allows true linear operation of the triode with a real load resistance. Unfortunately, this is strictly linear only for a constant load resistance, so depends on speaker impedance variation being small.
Don
"Have you made any measurements of swing?"
Yes, first off, one must get into the mindset of using current levels when using this circuit rather than voltage. Its not a suitable circuit for a high impedance voltage input stage, but more suited to intermediate or even output stages of an amplifier. So, first off, the input of an amplifier would be a voltage to current stage, at small input signal levels, before entering any current mirror stages.
The non-linear resistance on the input is not nearly as bad as the exponential non-linearity in SS current mirrors, so actual scope measurements on the input will show some voltage swing, in fact, this swing will obviously match the operating voltage swing of the pentode grid in a given design. The SS current mirrors are much more tricky to get accurate matching because of the exponential characteristic, so usually only appear in ICs where identical transistors are readily available. The V tube current mirror is actually more practical for discrete component design due to its more gradual 3/2 power law. (and a trimpot in the circuit to set grid bias level and a selectable number of diodes in series and a fair amount of patience on the designer's part!)
Since a constant current gain is enforced, the input current will be simply related to the output current by this gain factor.
I have gotten current gains up to around 97 using a 9KC6 pentode (+13V on G3, 100V on G2) and 6x series 6JU8 diode sections. Using 9006 diodes about doubles the gain yet versus the 6JU8 diodes. What voltage appears on the output of course depends on the load resistance (at least for a pentode with sufficiently high output resistance spec) times the current.
Higher current gain comes from using a smaller diode and higher gm pentode. Too small a diode however may get one into noise problems if the current draw is too small. The more series diodes one has to use to get the scaling correct, the higher the input (avg.) impedance will be and hence the input voltage swing for a given input current swing, but this is not a freely adjustable parameter.
When thinking in current gain mode, instead of voltage gain mode, one actually would prefer the input node to be low impedance so it doesn't swing around and affect the previous stage's output resistance. Only the last output stage reverts back to voltage swing into a load. So an "ideal" current mirror amplifier would see no voltage signals whatsoever internally (Notice that this KILLS Miller effects nicely!) , just on the output. Of course, having a pentode for output with very high output impedance is another problem, so usually one would put a cathode follower, or similar, on the output to lower output impedance or use negative feedback to lower Zout.
"diode(s)) to correct for the similarly non-linear curve of a tube"
Exactly. The diode is a 2/3 power law non-linear resistor and the pentode is a 3/2 power law non-linear Gm. The output is the 3/2 power function of the 2/3 power function of the input. For a function of a function, the exponents get multiplied (rather than added as for a multiplication function). More intuitively, for a given pentode grid voltage, the diode's input current will be the 3/2 power of the grid voltage and likewise the pentode's output current will be the 3/2 power of the grid voltage. So input to output current ratio is linear. The current gain comes simply from geometric scaling of the devices, use a bigger pentode (or higher gm) and you get more current out for the same grid voltage. Use a smaller diode and you get less input current.
This current mirror operation is incidentally quite similar to positive grid operation of a triode or pentode, everything just built into the same tube then. Only problem is the diode (grid) current subtracts from the plate current, eventually hogging the cathode current and causing saturation. So, not as linear as the current mirror configuration.
Another variation on this current mirror idea, (in fact it was invented this way I think) is to reverse the order. Pentode first, then a big diode for load impedance. This is the voltage mirror, and gives linear voltage gain. Unfortunately, its not too practical, as the load is the diode, nothing else allowed except maybe just a grid load. Not to mention that it takes a big diode to correct a small pentode. (clunk -- tossing into the dustbin)
Another useful application of the current mirror could be as a linearity corrector for a SET amplifier, the diode input corrector (with additional resistor) gets applied to the input of the triode. This allows true linear operation of the triode with a real load resistance. Unfortunately, this is strictly linear only for a constant load resistance, so depends on speaker impedance variation being small.
Don
I've been using SS current mirrors as loads on all my amps the last couple of years.
I dont know how EC8010 only got mV of swing with a ECC83 when I get hundreds of volts with every tube I've used so far. It's the SS mirror which is limiting the voltage swings since few P-channel MOSFETs or PNP bipolars go much higher(lower?) than -300V.
To me this topology is the easiest way to make amps with few stages.
I dont know how EC8010 only got mV of swing with a ECC83 when I get hundreds of volts with every tube I've used so far. It's the SS mirror which is limiting the voltage swings since few P-channel MOSFETs or PNP bipolars go much higher(lower?) than -300V.
To me this topology is the easiest way to make amps with few stages.
details details ...
Just a note, since the current mirror is a current to current gain device, it needs to be driven from a current source. So a typical amplifier circuit will consist of a conventional voltage to current tube stage (a pentode or pentode cascoded triode), but loaded by a current source or high value resistor, not a normal load resistor. Then capacitor coupled to the input of the subsequent current mirror stage. The same goes for mirror to mirror stage coupling. This might look a little crazy since a current source load on top of a current source driver would normally lead to undefined or rail pinned voltages, but the input diode impedance of the subsequent current mirror stage is what imposes order here, at least AC wise. With the capacitor coupling you still have an undefined load for the DC op. level of the current sources, so using a high value load resistor or a high L inductor load is called for. (Similarly, the diode input needs some steady DC current level too, to set the mirror stage DC operating point.)
But with current to current devices, another form of coupling is suitable. Using a current source load or high value resistor load to B+ for the V to current stage, the coupling to the mirror stage can be just a resistor from plate to diode input. This resistor is sized so that the idle current operating point (actually idle current of the driving tube minus idle current of the current source load) times the resistance gives the V drop down to the mirror stage diode/pentode grid level. (This residual current also sets the DC operating point for the diode and current mirror.) A capacitor CAN be placed across the coupling resistor for psychological comfort, but in theory at least, no loss of gain is caused by a coupling resistor in the series path of a current to current connection. The same current in, comes out the other end of the resistor.
This "current mode" thinking can be a little puzzling at times. For instance, to measure the current gain of a mirror stage on an X-Y scope display or dual channel scope .. or whatever, the input to the current mirror must be driven by a current source, not a voltage source. I usually just use a high voltage source with a high value resistor to the currrent mirror input. But remember, it is current you want to measure for linearity, not voltages. (Of course the output current is easily converted to voltage with just a conventional load resistor)
As the above resistive coupling technique hints, things can get complicated fast, since this provides DC coupling between stages.
Operating point drift can become a problem. So, unless you intend to make a DC coupled amplifier with a DC bias servo, I suggest some capacitor isolation or DC only feedback loops, or possibly a DC and AC global feedback loop. This is strange territory for usual tube design.
Don
Oops, Just saw this:
"I've been using SS current mirrors as loads on all my amps the last couple of years. ..."
The circuit most people are using for a current source load for tubes is actually a current source (or load) and not a current mirror in its intended configuration. The confusion of a current mirror load has arisen because current mirrors are often used to make a current source (or load) by programming them with a fixed input level. But just the output connection is being used for the connection to the plate, which is at high impedance with large voltage swing capability. The INPUT terminal for a SS current MIRROR is typically a low impedance with limited voltage swing. Using a SS current mirror input for a plate load connection will generally pretty much freeze the plate voltage, but will echo the current on its output. Very similar to a cascode stage (or grounded grid), except conveniently inverts the current output.
Hope I clarified things, rather than confusing them further. The English language is running out of terms to describe electronics well these days.
Just a note, since the current mirror is a current to current gain device, it needs to be driven from a current source. So a typical amplifier circuit will consist of a conventional voltage to current tube stage (a pentode or pentode cascoded triode), but loaded by a current source or high value resistor, not a normal load resistor. Then capacitor coupled to the input of the subsequent current mirror stage. The same goes for mirror to mirror stage coupling. This might look a little crazy since a current source load on top of a current source driver would normally lead to undefined or rail pinned voltages, but the input diode impedance of the subsequent current mirror stage is what imposes order here, at least AC wise. With the capacitor coupling you still have an undefined load for the DC op. level of the current sources, so using a high value load resistor or a high L inductor load is called for. (Similarly, the diode input needs some steady DC current level too, to set the mirror stage DC operating point.)
But with current to current devices, another form of coupling is suitable. Using a current source load or high value resistor load to B+ for the V to current stage, the coupling to the mirror stage can be just a resistor from plate to diode input. This resistor is sized so that the idle current operating point (actually idle current of the driving tube minus idle current of the current source load) times the resistance gives the V drop down to the mirror stage diode/pentode grid level. (This residual current also sets the DC operating point for the diode and current mirror.) A capacitor CAN be placed across the coupling resistor for psychological comfort, but in theory at least, no loss of gain is caused by a coupling resistor in the series path of a current to current connection. The same current in, comes out the other end of the resistor.
This "current mode" thinking can be a little puzzling at times. For instance, to measure the current gain of a mirror stage on an X-Y scope display or dual channel scope .. or whatever, the input to the current mirror must be driven by a current source, not a voltage source. I usually just use a high voltage source with a high value resistor to the currrent mirror input. But remember, it is current you want to measure for linearity, not voltages. (Of course the output current is easily converted to voltage with just a conventional load resistor)
As the above resistive coupling technique hints, things can get complicated fast, since this provides DC coupling between stages.
Operating point drift can become a problem. So, unless you intend to make a DC coupled amplifier with a DC bias servo, I suggest some capacitor isolation or DC only feedback loops, or possibly a DC and AC global feedback loop. This is strange territory for usual tube design.
Don
Oops, Just saw this:
"I've been using SS current mirrors as loads on all my amps the last couple of years. ..."
The circuit most people are using for a current source load for tubes is actually a current source (or load) and not a current mirror in its intended configuration. The confusion of a current mirror load has arisen because current mirrors are often used to make a current source (or load) by programming them with a fixed input level. But just the output connection is being used for the connection to the plate, which is at high impedance with large voltage swing capability. The INPUT terminal for a SS current MIRROR is typically a low impedance with limited voltage swing. Using a SS current mirror input for a plate load connection will generally pretty much freeze the plate voltage, but will echo the current on its output. Very similar to a cascode stage (or grounded grid), except conveniently inverts the current output.
Hope I clarified things, rather than confusing them further. The English language is running out of terms to describe electronics well these days.
6L6 output current mirror
Hello Smoking amp, this thread has got me all excited, it seems perfect for what I have been trying to do, a current drive amp for my Lowther speakers.
What I'm thinking about is two stages of BDTs (yes I'm still doing that) as transconductance amp (its very good at this) driving a 6L6 SE parafeed output stage.
I have everything pretty much figured out except the diodes to use. I have all the parts except the diodes. I was wondering if I could use a 12AX7 as a diode here. Looking at the curves with the grid at -.5V it looks like a pretty good match to the 6L6 curve, I didn't understand your post about why this won't work. Could you explain that in a little more detail?
Obviously the biasing here gets a little hairy, both the diode and the pentode grid have to be on the right part of the curve from an AC standpoint a well as being in the right place DC wise. A little tricky but not insurmountable.
I presume this is a case of working things out using the published curves, building it, trying it out and finding the you weren't even close and have to tweak things a lot to get the curves to match
I started looking at a PP 6L6 output stage and what it would take for everything to come out right and balanced and gave up, I'll start with a nice "simple" SE stage first.
I don't know if the 6L6 is a good candidate for this or not, but I have a bunch (my BDT amp was originally going to be BDT as phase splitter for 6L6 PP amp so I have a lot of parts for this already)
Thanks for bringing this up, it seems so obvious now, but I never would have thought of it.
John S.
Hello Smoking amp, this thread has got me all excited, it seems perfect for what I have been trying to do, a current drive amp for my Lowther speakers.
What I'm thinking about is two stages of BDTs (yes I'm still doing that) as transconductance amp (its very good at this) driving a 6L6 SE parafeed output stage.
I have everything pretty much figured out except the diodes to use. I have all the parts except the diodes. I was wondering if I could use a 12AX7 as a diode here. Looking at the curves with the grid at -.5V it looks like a pretty good match to the 6L6 curve, I didn't understand your post about why this won't work. Could you explain that in a little more detail?
Obviously the biasing here gets a little hairy, both the diode and the pentode grid have to be on the right part of the curve from an AC standpoint a well as being in the right place DC wise. A little tricky but not insurmountable.
I presume this is a case of working things out using the published curves, building it, trying it out and finding the you weren't even close and have to tweak things a lot to get the curves to match
I started looking at a PP 6L6 output stage and what it would take for everything to come out right and balanced and gave up, I'll start with a nice "simple" SE stage first.
I don't know if the 6L6 is a good candidate for this or not, but I have a bunch (my BDT amp was originally going to be BDT as phase splitter for 6L6 PP amp so I have a lot of parts for this already)
Thanks for bringing this up, it seems so obvious now, but I never would have thought of it.
John S.
Hi John,
First, its good to hear the BDT idea is still going. I have been thinking of using one for a first input stage to an amp also.
The 12AX7 may work OK for a diode string sub since you are using it for a class A SE stage where it will be biased up into the middle of its operating range. Worth a try anyway. The problem I have found with using a triode as a diode element is at low currents where the triode's grid causes some shadowing of the cathode, so that different parts of the cathode are operating with different electric field strength. (so called "island effect" or inselbildung sp?) Causing some distortion of the 3/2 s power curve at low currents. Generally, the larger the voltage swing required on the driven pentode grid, the more series diodes needed. So being able to use a triode with its adjustable geometry factor via its grid bias would be a nice economizing of parts. Rig up the scope for X-Y display and tweak away with variable bias voltages for the 12AX7 grid-cathode bias, and the bias #2 from 6L6 cathode to 12AX7 cathode. Remember, it is input current that is linear with the output current. Good luck!
Don
First, its good to hear the BDT idea is still going. I have been thinking of using one for a first input stage to an amp also.
The 12AX7 may work OK for a diode string sub since you are using it for a class A SE stage where it will be biased up into the middle of its operating range. Worth a try anyway. The problem I have found with using a triode as a diode element is at low currents where the triode's grid causes some shadowing of the cathode, so that different parts of the cathode are operating with different electric field strength. (so called "island effect" or inselbildung sp?) Causing some distortion of the 3/2 s power curve at low currents. Generally, the larger the voltage swing required on the driven pentode grid, the more series diodes needed. So being able to use a triode with its adjustable geometry factor via its grid bias would be a nice economizing of parts. Rig up the scope for X-Y display and tweak away with variable bias voltages for the 12AX7 grid-cathode bias, and the bias #2 from 6L6 cathode to 12AX7 cathode. Remember, it is input current that is linear with the output current. Good luck!
Don
a few more thoughts
If you can't get the 12AX7 to work satisfactorily as a diode corrector, the other approach, which is what I am using, is to use a separate output voltage gain stage with high Gm, followed by a partial cathode follower output. A high Gm pentode generally only requires a couple of 6JU8 multi-diode tubes for its linear corrector.
My present plan for a P-P class AB amp. using this is as follows: LTP or BDT input stage, followed by 6LE8 or 9KC6 (or 12HG7, 12GN7 ... for more conventional minded) high Gm pentodes with 6JU8 correctors. Then a circlotron like (search on "elliptron" and "Hawksford" in tube section ) output stage (6HJ5) with Hawksford style error corrector (6J11 maybe). The current mode output of the VT current mirror is a natural fit to the summing node requirements of the Hawksford style output error corrector.
Then, maybe optionally, a global feedback loop from the xfmr output to the LTP/BDT. Since this amp will have a BUNCH of trimpots for setting all the corrector linearities, I will also include a SS diff. amp. across the LTP/BDT inputs for a Baxandall style distortion monitor which can be connected to a scope or headphones while tweeking pot settings. Finally, the global feedback loop will have provision for using a REVERSE connected constant current triode stage as the feedback network attenuator (in place of the usual resistor attenuator network), this will allow simulation of single triode SE operation since the triode sets the overall gain rule.
Before I can get back to work on this monster project however, I need to solve a monster math problem for another project. Any math geniuses out there familiar with finding conjugate harmonic functions for a given toroidal harmonic function? Help! (They are worse than even the dreaded hypergeometric functions as far as I can tell!)
Don
If you can't get the 12AX7 to work satisfactorily as a diode corrector, the other approach, which is what I am using, is to use a separate output voltage gain stage with high Gm, followed by a partial cathode follower output. A high Gm pentode generally only requires a couple of 6JU8 multi-diode tubes for its linear corrector.
My present plan for a P-P class AB amp. using this is as follows: LTP or BDT input stage, followed by 6LE8 or 9KC6 (or 12HG7, 12GN7 ... for more conventional minded) high Gm pentodes with 6JU8 correctors. Then a circlotron like (search on "elliptron" and "Hawksford" in tube section ) output stage (6HJ5) with Hawksford style error corrector (6J11 maybe). The current mode output of the VT current mirror is a natural fit to the summing node requirements of the Hawksford style output error corrector.
Then, maybe optionally, a global feedback loop from the xfmr output to the LTP/BDT. Since this amp will have a BUNCH of trimpots for setting all the corrector linearities, I will also include a SS diff. amp. across the LTP/BDT inputs for a Baxandall style distortion monitor which can be connected to a scope or headphones while tweeking pot settings. Finally, the global feedback loop will have provision for using a REVERSE connected constant current triode stage as the feedback network attenuator (in place of the usual resistor attenuator network), this will allow simulation of single triode SE operation since the triode sets the overall gain rule.
Before I can get back to work on this monster project however, I need to solve a monster math problem for another project. Any math geniuses out there familiar with finding conjugate harmonic functions for a given toroidal harmonic function? Help! (They are worse than even the dreaded hypergeometric functions as far as I can tell!)
Don
One other quick question, what is R2 for? It seems like its purpose is to generate negative feedback, but why do you want to do that if you are already linearizing the transfer function? It would also cut down on the gain which is something I don't need to be happening with the 6L6.
Thanks,
John S.
Thanks,
John S.
R1 and R2
The R1 and R2 resistors are used like normal degeneration resistors to improve linearity if the diode does not perfectly match the pentode. In this case, however, no gain is lost if the resistors are sized so as to match the original nominal current gain of the diode and pentode. For a perfectly matched diode and pentode, the resistors are indeed redundant. They will just cost a little voltage headroom is all. The degeneration idea is used in SS current mirrors since they are more difficult to match identically with their exponential characteristics.
For a triode gain stage, instead of a pentode stage, the resistor in series with the diodes can be made a little higher in value to effectively cancel out the internal voltage feedback from the plate (as long as the plate load is a simple fixed resistance.) (plate voltage swing divided by Mu is the extra voltage swing needed on the triode grid for that input current swing to cancel out the plate effect)
Another thought on getting a high power current output stage:
could use a high Gm pentode in the VT current mirror, then a cascode stage above it for high voltage swing.
Finally, a far out idea to try: The BDTs have internal voltage feedback from their plates to the beam deflection besides the deflectors. This effect sets a max gain limit to the BDTs. Not sure whether the plate effect will be 3/2 power or linear like the deflectors. If it is in fact 3/2 power, might be an elegant way to get an adjustable geometry factor diode equivalent without grid island effect. Adjusting a bias voltage across the deflectors would adjust the geometry factor then. Probably a linear plate effect though. Another tube with some possiblility as a variable diode could be that nutty gated pentode 6BN6.
Oh, and another small note, the screen voltage on the pentode in a VT current mirror also has effects on the geometry factor, so if you can't get the diode geometry factor quite right, can adjust the pentode screen voltage.
Don B.
The R1 and R2 resistors are used like normal degeneration resistors to improve linearity if the diode does not perfectly match the pentode. In this case, however, no gain is lost if the resistors are sized so as to match the original nominal current gain of the diode and pentode. For a perfectly matched diode and pentode, the resistors are indeed redundant. They will just cost a little voltage headroom is all. The degeneration idea is used in SS current mirrors since they are more difficult to match identically with their exponential characteristics.
For a triode gain stage, instead of a pentode stage, the resistor in series with the diodes can be made a little higher in value to effectively cancel out the internal voltage feedback from the plate (as long as the plate load is a simple fixed resistance.) (plate voltage swing divided by Mu is the extra voltage swing needed on the triode grid for that input current swing to cancel out the plate effect)
Another thought on getting a high power current output stage:
could use a high Gm pentode in the VT current mirror, then a cascode stage above it for high voltage swing.
Finally, a far out idea to try: The BDTs have internal voltage feedback from their plates to the beam deflection besides the deflectors. This effect sets a max gain limit to the BDTs. Not sure whether the plate effect will be 3/2 power or linear like the deflectors. If it is in fact 3/2 power, might be an elegant way to get an adjustable geometry factor diode equivalent without grid island effect. Adjusting a bias voltage across the deflectors would adjust the geometry factor then. Probably a linear plate effect though. Another tube with some possiblility as a variable diode could be that nutty gated pentode 6BN6.
Oh, and another small note, the screen voltage on the pentode in a VT current mirror also has effects on the geometry factor, so if you can't get the diode geometry factor quite right, can adjust the pentode screen voltage.
Don B.
Triodes as adjustable Diodes - revised
After some more thinking about the shape of the current transfer curve using a fixed bias triode as the corrector diode, the problem may NOT be from "island effect". At least not primarily. The curve is linear except at low current, where it sort of cuts off at a minimum threshold current (instead of zero current). But the plate to cathode voltage IS near zero at this cutoff. Since I couldn't adjust it out with the "diode" cathode to pentode cathode bias supply, I too quickly assumed it must be a grid "island effect" when it actually fits just a simple cathode emission cutoff model. There are some squiggly bends in the curve as well as the cutoff, so some "island effect" may be present too.
What I think is happening is the grid voltage defines a new virtual cathode for the "diode", so the linear current transfer curve has a zero current intercept at the plate voltage equal to the negative grid voltage, but actual cathode emission cuts off a little earlier when plate to cathode V is zero.
If this is indeed the case, then the way to get a full range adjustable diode corrector using a triode leads to a different sort of triode than usual. The usual triodes I have tried for correctors have too much conductivity, or equiv. too close a cathode to plate spacing, and one needs to put a negative bias on the grid to adjust conductivity down, leading to the cutoff problem at low current. Instead, what is needed is a triode with too low a conductivity, or equiv. too great a cathode to plate spacing. Fortunately, such a critter exists. The long forgotten beam triode voltage regulator tubes should fit the need. One would use a small positive fixed bias on the grid. (at least I hope this would work to adjust cond. up) Unfortunately, they are a bit bulky in size. All the ones I am aware of are in the 25 to 30 Watt plate dissipation range. At least they are throw away cheap. I'll give these a try in the V T current mirror circuit when I get back from vacation. (Have some laying around anyway)
(Note: just using a physically small triode to get lower conductivity probably won't work since the close plate to cathode spacing will cause it to operate on the wrong part of the 3/2 power curve relative to the pentode)
Since the beam triode tubes are kinda big, they could also be used in the VT "voltage mirror" circuit too. Here, they act as the compensating non-linear load for a pentode. Unfortunately, only the diode load is allowed for correction, any resistance destroys the correction effect. But still useable for driving a subsequent grid.
Don B.
After some more thinking about the shape of the current transfer curve using a fixed bias triode as the corrector diode, the problem may NOT be from "island effect". At least not primarily. The curve is linear except at low current, where it sort of cuts off at a minimum threshold current (instead of zero current). But the plate to cathode voltage IS near zero at this cutoff. Since I couldn't adjust it out with the "diode" cathode to pentode cathode bias supply, I too quickly assumed it must be a grid "island effect" when it actually fits just a simple cathode emission cutoff model. There are some squiggly bends in the curve as well as the cutoff, so some "island effect" may be present too.
What I think is happening is the grid voltage defines a new virtual cathode for the "diode", so the linear current transfer curve has a zero current intercept at the plate voltage equal to the negative grid voltage, but actual cathode emission cuts off a little earlier when plate to cathode V is zero.
If this is indeed the case, then the way to get a full range adjustable diode corrector using a triode leads to a different sort of triode than usual. The usual triodes I have tried for correctors have too much conductivity, or equiv. too close a cathode to plate spacing, and one needs to put a negative bias on the grid to adjust conductivity down, leading to the cutoff problem at low current. Instead, what is needed is a triode with too low a conductivity, or equiv. too great a cathode to plate spacing. Fortunately, such a critter exists. The long forgotten beam triode voltage regulator tubes should fit the need. One would use a small positive fixed bias on the grid. (at least I hope this would work to adjust cond. up) Unfortunately, they are a bit bulky in size. All the ones I am aware of are in the 25 to 30 Watt plate dissipation range. At least they are throw away cheap. I'll give these a try in the V T current mirror circuit when I get back from vacation. (Have some laying around anyway)
(Note: just using a physically small triode to get lower conductivity probably won't work since the close plate to cathode spacing will cause it to operate on the wrong part of the 3/2 power curve relative to the pentode)
Since the beam triode tubes are kinda big, they could also be used in the VT "voltage mirror" circuit too. Here, they act as the compensating non-linear load for a pentode. Unfortunately, only the diode load is allowed for correction, any resistance destroys the correction effect. But still useable for driving a subsequent grid.
Don B.
Hi Don,
what I was thinking of doing was putting a plate choke on the BDT, then direct couple to the plate of the triode. That runs the triode in a nice high voltage/current spot in its curve. Then cap couple the triode to the pentode.
Today I was looking at the ECL85 which has a triode and pentode in one envelope, it looks like if I run the BDT at a little lower voltage than I normally do (150-200V) I can run the triode section right where I want it to be on its curve with what looks like 2-3 mA.
It looks like a really stiff screen is needed for this use, maybe a voltage regulator. I have a nice little ecl85 series regulator from Frank (which is why I have the ecl85s on hand!) which would make a good adjustable screen regulator.
With the ecl85s I might even get a PP version up and running. It would make a nice little 3 watt current source amp with 1 BDT and 2 ecl85s per channel. Not bad.
I think I'll try the SE version first though.
John S.
what I was thinking of doing was putting a plate choke on the BDT, then direct couple to the plate of the triode. That runs the triode in a nice high voltage/current spot in its curve. Then cap couple the triode to the pentode.
Today I was looking at the ECL85 which has a triode and pentode in one envelope, it looks like if I run the BDT at a little lower voltage than I normally do (150-200V) I can run the triode section right where I want it to be on its curve with what looks like 2-3 mA.
It looks like a really stiff screen is needed for this use, maybe a voltage regulator. I have a nice little ecl85 series regulator from Frank (which is why I have the ecl85s on hand!) which would make a good adjustable screen regulator.
With the ecl85s I might even get a PP version up and running. It would make a nice little 3 watt current source amp with 1 BDT and 2 ecl85s per channel. Not bad.
I think I'll try the SE version first though.
John S.
Hi John,
Running the triode at high V/high current may get the curves to match up with the pentode, but you may get very little current gain, if any, that way. So better check the resultant gain.
I don't have any tube books with me here on vacation, anyone know of any small, hopefully 7 or 9 pin, version of the beam triode voltage regulators? Or anything vaguely similar?
Don B.
Running the triode at high V/high current may get the curves to match up with the pentode, but you may get very little current gain, if any, that way. So better check the resultant gain.
I don't have any tube books with me here on vacation, anyone know of any small, hopefully 7 or 9 pin, version of the beam triode voltage regulators? Or anything vaguely similar?
Don B.
Hi Don,
well I finally built the thing. I tried an ECL85 using the triode as the diode.
For the first attempt I took the plate from the BDT directly to the plate of the triode, then cap coupled that tp the grid of the pentode with fixed bias from 9v batteries and pots.
I first looked at the pentode driven directly, very non linear looking waveform (using a triangle wave as source). With the triode in the circuit I got some correction but not much no matter what I used as triode grid voltage.
I then decided to hook it up as you originally speced with the BDT cap coupled to the triaode plate with the bias on the cathode. Turned it on and looked first at the triode plate, woa, a very distorted waveform! I then looked at the pentode plate and it was a perfect triangle wave, it worked!!
Then I remembered that you didn't think it would work very well, so I started lto try and find out why it was working so well. Then I realized I had left the triode grid hooked up to -4 v even though the cathode was at -20v, I was running the grid at +16v with respect to the cathode!! I thought there must be a lot of grid current and I didn't want anything exploding on me, so I unplugged it and went to bed.
The next day I measured it and found I was putting 15mA into the grid, probably not a good thing for a small low power signal triode.
So I tried hooking he pot for the triode grid up to the same three batteries I was using for the pentode bias and tried setting the triode grid just a little positive but it would not let me. I was gettig some weird feedback through the bias pots, adjusting the triode grid pot changed the current through the cathode which changed the grid to cathode voltage etc. The result was I could never get the grid positive with respect to the cathode. Tonight I'm building some voltage regulators so I hopefully can set the grid positive by just a few volts and see if that still gives a linear response.
That beutiful perfectly straight sidded triangle wave coming out of a pentode with no feedback was an amazing sight. I hope it sounds good too!
Thanks for bringing this whole concept up.
John S.
well I finally built the thing. I tried an ECL85 using the triode as the diode.
For the first attempt I took the plate from the BDT directly to the plate of the triode, then cap coupled that tp the grid of the pentode with fixed bias from 9v batteries and pots.
I first looked at the pentode driven directly, very non linear looking waveform (using a triangle wave as source). With the triode in the circuit I got some correction but not much no matter what I used as triode grid voltage.
I then decided to hook it up as you originally speced with the BDT cap coupled to the triaode plate with the bias on the cathode. Turned it on and looked first at the triode plate, woa, a very distorted waveform! I then looked at the pentode plate and it was a perfect triangle wave, it worked!!
Then I remembered that you didn't think it would work very well, so I started lto try and find out why it was working so well. Then I realized I had left the triode grid hooked up to -4 v even though the cathode was at -20v, I was running the grid at +16v with respect to the cathode!! I thought there must be a lot of grid current and I didn't want anything exploding on me, so I unplugged it and went to bed.
The next day I measured it and found I was putting 15mA into the grid, probably not a good thing for a small low power signal triode.
So I tried hooking he pot for the triode grid up to the same three batteries I was using for the pentode bias and tried setting the triode grid just a little positive but it would not let me. I was gettig some weird feedback through the bias pots, adjusting the triode grid pot changed the current through the cathode which changed the grid to cathode voltage etc. The result was I could never get the grid positive with respect to the cathode. Tonight I'm building some voltage regulators so I hopefully can set the grid positive by just a few volts and see if that still gives a linear response.
That beutiful perfectly straight sidded triangle wave coming out of a pentode with no feedback was an amazing sight. I hope it sounds good too!
Thanks for bringing this whole concept up.
John S.
Triodes as diode correctors
Hi John,
Good to hear you got something working. From your description of using a cap. in series with the triode plate, I assume you have some hi value pullup resistor on the triode plate as well to provide DC bias current for the "Diode" so the cap. doesn't just charge up until the "diode" stops conducting.
You should be able to dispense with the cap. eventually by just using a high value resistor from the BDT plate down to the "Diode" plate and pentode grid connection. Have to figure the idle current for the diode and then the voltage drop from the BDT plate down to the "Diode" for calculating the resistor value. But, for testing/design its nice to have the freedom to change diode (or triode) idle current given by the cap. isolation. (Note, a high value coupling resistor doesn't cause any signal attenuation in a current coupled amplifier, same current in = same current out, just voltage drop.)
I did some testing here using some beam triode tubes for the diode corrector. I found they do indeed need some positive grid voltage (with respect to their cathode) to operate as correctors. About +1 to 4 Volts. I was surprised to find however that the low current cutoff problem that I found earlier using normal triodes was not fixed this way. The beam triodes also seem to like a little positive voltage on their beam electrode as well, around +2V to +15V or so.
I went back and tried a 12AX7 and 5687 triode, and they operate well with positive grid volts too, but still with the low current cutoff. About +1 to 2 Volts on the grid. What I found is that the higher the positive grid volts on the triode, the lower the cutoff current threshold for the stage, but also the lower the current gain for the stage. (one has to re-adjust the DC bias on the "diode" cathode with respect to the pentode cathode too as one changes the triode grid bias of course) (I was using a 9KC6 for the pentode with +13V on G3, +100V on G2 for all tests) So it seems there is no advantage in using the bulky size beam triode tubes. Also, I noticed that there is a zone in the transition from negative to positive grid bias where the stage transfer curve kinks up badly, so one has to go with one or the other only.
I wish I could figure out some way to get rid of that low current cutoff problem with triodes used as "diodes" for use in class AB stages, but one can still just operate up in the middle of the linear current range for class A.
Using normal triodes (for "diode" correctors), one will get higher stage current gains with higher Rp (plate resistance) triodes in general, although this is obviously a parameter that varies with current (as well it should in order to act as a 2/3 power corrector).
I notice that the triode in the ECL85 is rated at around 10K Rp. You will get better current gain using something more like a 12AX7 triode with its 100K Rp.
I will have to do a little searching to see if their are any really high Rp triodes around. I did try a 6BK4 HV triode which has really-REALLY! high Rp, but it doesn't work. Like an open circuit with low voltage on its plate, even with a positive grid. Only other weird tube I can think of off hand to try next is that 6BN6 gate tube. Should have a fair amount of distance between cathode and plate and a small cathode emission area. Will obviously require some playing around with multiple bias voltages on all the other extraneous elements to see if it will act as a diode corrector. Hmmm, maybe a BDT tube might work too, lots of distance between the cathode and plates there. Connecting the two BDT plates together will probably get back to a 2/3 power law resistor.
Don
Hi John,
Good to hear you got something working. From your description of using a cap. in series with the triode plate, I assume you have some hi value pullup resistor on the triode plate as well to provide DC bias current for the "Diode" so the cap. doesn't just charge up until the "diode" stops conducting.
You should be able to dispense with the cap. eventually by just using a high value resistor from the BDT plate down to the "Diode" plate and pentode grid connection. Have to figure the idle current for the diode and then the voltage drop from the BDT plate down to the "Diode" for calculating the resistor value. But, for testing/design its nice to have the freedom to change diode (or triode) idle current given by the cap. isolation. (Note, a high value coupling resistor doesn't cause any signal attenuation in a current coupled amplifier, same current in = same current out, just voltage drop.)
I did some testing here using some beam triode tubes for the diode corrector. I found they do indeed need some positive grid voltage (with respect to their cathode) to operate as correctors. About +1 to 4 Volts. I was surprised to find however that the low current cutoff problem that I found earlier using normal triodes was not fixed this way. The beam triodes also seem to like a little positive voltage on their beam electrode as well, around +2V to +15V or so.
I went back and tried a 12AX7 and 5687 triode, and they operate well with positive grid volts too, but still with the low current cutoff. About +1 to 2 Volts on the grid. What I found is that the higher the positive grid volts on the triode, the lower the cutoff current threshold for the stage, but also the lower the current gain for the stage. (one has to re-adjust the DC bias on the "diode" cathode with respect to the pentode cathode too as one changes the triode grid bias of course) (I was using a 9KC6 for the pentode with +13V on G3, +100V on G2 for all tests) So it seems there is no advantage in using the bulky size beam triode tubes. Also, I noticed that there is a zone in the transition from negative to positive grid bias where the stage transfer curve kinks up badly, so one has to go with one or the other only.
I wish I could figure out some way to get rid of that low current cutoff problem with triodes used as "diodes" for use in class AB stages, but one can still just operate up in the middle of the linear current range for class A.
Using normal triodes (for "diode" correctors), one will get higher stage current gains with higher Rp (plate resistance) triodes in general, although this is obviously a parameter that varies with current (as well it should in order to act as a 2/3 power corrector).
I notice that the triode in the ECL85 is rated at around 10K Rp. You will get better current gain using something more like a 12AX7 triode with its 100K Rp.
I will have to do a little searching to see if their are any really high Rp triodes around. I did try a 6BK4 HV triode which has really-REALLY! high Rp, but it doesn't work. Like an open circuit with low voltage on its plate, even with a positive grid. Only other weird tube I can think of off hand to try next is that 6BN6 gate tube. Should have a fair amount of distance between cathode and plate and a small cathode emission area. Will obviously require some playing around with multiple bias voltages on all the other extraneous elements to see if it will act as a diode corrector. Hmmm, maybe a BDT tube might work too, lots of distance between the cathode and plates there. Connecting the two BDT plates together will probably get back to a 2/3 power law resistor.
Don
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