Is the CFB topology superior, and why? - Page 11 - diyAudio
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Old 29th September 2012, 01:24 AM   #101
Bonsai is offline Bonsai  Taiwan
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Quote:
Originally Posted by rcw666 View Post
The amplifier bandwidth is limited by the current available to slew the compensation capacitor.

In an amplifier with no access to the compensation capacitor that is stabilised for unity gain the bandwidth is limited by the value of this capacitor, and in the voltage feedback scheme the current available to slew this capacitor is fixed.

In the CFB topology however the feedback resistor sets the current available to slew this capacitor, thus over a given range the bandwidth is only dependent upon this resistor, this is the Go term in the expression that I posted, Zt is constant for a given Cc provided that Rf is also constant, this is true over the range specified in the data sheets.

Your modeled circuits do not have the pole splitting capacitor, so I am assuming that the slew is limited by the transistor characteristics alone, putting the capacitor in the simulation might be informative.
rcw
Just coming back to this comment, I think your comment about the slewing current being set by the feedback resistor in CFA is the key insight, and this why it's the value of the feedback resistor that's often recommended as the primary method of comp for this topology. Because the output buffer is sourcing this current (and the feedback resistor can therefore be low in value), you can have very high slewing currents. Typically a VFB might have 10mA available but you can configure a CFB easily for 5x or 10x this figure.
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Old 29th September 2012, 06:07 AM   #102
catalin is offline catalin  Romania
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Quote:
Originally Posted by rcw666 View Post
Leach pointed out long ago that there is no bandwidth advantage in using a cascode in a feedback amplifier, it still needs a compensation capacitor that splits the poles and puts one high enough to ensure stability.
rcw
Leach was saying that the cascode doesn't add bennefits if it is positioned on both signals,the input and the feedback.
But if the input has 2 poles ,one at low and the feedback signal just one because of common base stage then you have a shift of 90 degrees at high F not at low like in VFB .
in vfb you have a pole at low because the fb signals enters in differential pair in base so we have a common emitter stage which introduce a pole with -20db/decade. and for phase a -90 degree.in cfb the input for the fb is the emitter and there is a common base stage but this stage does not have a pole at low because there is no -180 degrees phase shift like common emitter has. in common base stage the miller parasitic capacity is the output capacity but in common emitter the capacity seen at the output is multiplied by the gain of the stage.please review the transistor gain stages.
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Old 29th September 2012, 10:16 AM   #103
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Quote:
Originally Posted by catalin View Post
"First, your frequency analysis measures the open loop gain under forced VFB conditions (and extreme ones)" Can you explain ?
In a CFB, the impedance of the FB network is an essential part of the amplifier.
If you remove it by driving the - node at zero impedance, you make the measurement in a way that is totally disconnected from the reality, where the FB network sets the loop gain, slew rate etc.
OK, it is the "true" open loop gain, but it is purely virtual, and it is completely useless for stability analysis for example. This amplifier is unstable.

Quote:
Elvee please excuse but your explanations are wrong .
Firstly you analyze the OLG with the AC signal between the output and the negative feedback .The reference for the negative input must be the GND beacuse we transform the voltage in current .If the negative reaction is in current then we should add a current generator but spice don't have such thing and then we shoul transform the V in I .
That is the proper way to analyze the loop gain (not totally accurate mathematically, but close enough since Zout is small compared to the FB network).
See Keantoken posts for example for more details

Quote:
You missplaced wrong the current source for the differentail input in Vfb .This will do other impedances in the nodes and will give small AC modified behaviour which is wrong .
Who said a VFB amplifier must have a classical, non-degenerated LTP at the input?
The amplifier has been converted to a VFB one, the (-) input impedance has been raised by a Hfe factor, and the impedance of the FB network has no more influence on the behavior of the amplifier (within reasonable limits).
You could convert the schematic to a more familiar looking one by splitting the 180 ohm into two 90 ohm, but this changes nothing.
The emitter follower could also be replaced by a diamond buffer, thus freeing the amplifier completely from bias current constraints. It would still remain a VFB.
Now the gain is set by the degeneration resistor, R2.
That is basically what I want to show: a CFB amplifier is essentially a degenerated amplifier, it is like adding emitter resistors in the input LTP; additionaly, the tail current can be split and manipulated to remove slew rate constraints

Quote:
Also I can show you a real amp CFB schematic has linear phase (0 degrees until 1 MHZ) who has also good phase reserve but you can't show me a VFB with the same phase because it doesn't exist .The theory says so !
You need to compensate it !
Always a Vfb will have one pole at very low F ,even there are no discrete capacitors .But the parasitic/internal Miller capacitor will give the look of my OLG VFB picture . If you use the cascode stage like CFB has, than Miller is not matter anymore .
Can you show a vfb compensated with linear phase ?I guess not ..
The amplifier you show is not stable without compensation.
0° phase at 1MHz means nothing: what counts is the variation of the group delay as a function of the frequency. We see on the plots that the group delay at 1MHz is indeed very small, but at medium frequencies it increases to 1.5µs. That behavior is worst than an amplifier having a flat group delay of 1.5µs from DC to 1MHz

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The last 2 pictures are the same . Group delays will not be the same when you have a compensated VFB .
I mixed up the pics. Here is the correct one.
The VFB version does not need more compensation than its CFB counterpart, because it has been designed to mimic exactly its behavior.
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Old 29th September 2012, 11:52 AM   #104
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Elvee, can you show VFB that is phase linear to at least 1mhz? Now that would convince a few people.
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Old 29th September 2012, 12:47 PM   #105
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Quote:
Originally Posted by catalin View Post
... in vfb you have a pole at low because the fb signals enters in differential pair in base so we have a common emitter stage which introduce a pole with -20db/decade
The most important question for this thread is: Why someone would place extra unlinear gain part in feedback path apart from resistors?, since the feedback current comes from the low Z output, so there's a plenty current to use. Intentionally putting unlinear gain part to FB path brings only bad stuff to the signal and correction. Plain logic. CFB OLG phase graph expalins it all.
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Old 29th September 2012, 02:40 PM   #106
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Quote:
Originally Posted by Lazy Cat View Post
The most important question for this thread is: Why someone would place extra unlinear gain part in feedback path apart from resistors?, since the feedback current comes from the low Z output, so there's a plenty current to use. Intentionally putting unlinear gain part to FB path brings only bad stuff to the signal and correction. Plain logic. CFB OLG phase graph explains it all.
Observation: I have noted that perceived soundstage size in monophonic changes size depending on feedback current. That's not the only way to do it, but the tallest order for a hi-fi grade audio amp needs all the help it can get. I wish this were easier to measure.

Simple example with a simple part--LM1875 non-inverting:
180k feedback resistor--bankrupt sound stage smashed flat and drawing attention directly to the speaker
100k feedback resistor--great improvement and lovely tone--soundstage is weirdly hallway shape (deep but not wide)
10k feedback resistor--perfect (equally wide and deep) soundstage and improved dynamics, but poor tone caveats.
Except for nesting/composite, there wasn't a solid answer.
However, AndrewT's even balance answer was 47k and with a low gain setting (a computer sound chip at max would not clip the 25 watt amp).
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Last edited by danielwritesbac; 29th September 2012 at 03:01 PM.
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Old 29th September 2012, 05:52 PM   #107
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Quote:
Originally Posted by Lazy Cat View Post
The most important question for this thread is: Why someone would place extra unlinear gain part in feedback path apart from resistors?, since the feedback current comes from the low Z output, so there's a plenty current to use. Intentionally putting unlinear gain part to FB path brings only bad stuff to the signal and correction. Plain logic. CFB OLG phase graph expalins it all.
Lazy Cat

I'm still not getting this, can you or Catalin expand on what you mean by an additional nonlinear stage?
Fundamentally isn't the error current developed the same and subject to the same transistor characteristics? The difference being only that vfb has a limited current?

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Old 29th September 2012, 06:17 PM   #108
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Elvee, I didn't say that those pictures belong to this schematic .
So if you want to see a good CFP schematic with phase reserve I can show you .
The pictures was only to present the diferences between the topologies .
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Old 29th September 2012, 08:36 PM   #109
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I am well aware Catalin that all active devices have an input and output time constant, this is precisely why dominant pole compensation is used.

As Leach also showed the best place for this is the stage in a feedback amplifier that has the most voltage gain.

Dominant pole compensation in effect increases the input time constant of this stage and decreases the output one, the pole splitting effect, giving the closed loop a first order convergence.

In this all other device time constants are at a sufficiently high frequency as not to matter, and at very high frequencies local pole zero cancellation is often used.

Your arguments regarding the input ltp time constants are irrelevant because the dominant pole and its split pole are the only thing that matters in the pass band, and these are virtually identical for a given band width in both vfb and cfb schemes.
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Old 29th September 2012, 11:21 PM   #110
catalin is offline catalin  Romania
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Originally Posted by rcw666 View Post
I am well aware Catalin that all active devices have an input and output time constant, this is precisely why dominant pole compensation is used.

As Leach also showed the best place for this is the stage in a feedback amplifier that has the most voltage gain.

Dominant pole compensation in effect increases the input time constant of this stage and decreases the output one, the pole splitting effect, giving the closed loop a first order convergence.

In this all other device time constants are at a sufficiently high frequency as not to matter, and at very high frequencies local pole zero cancellation is often used.

Your arguments regarding the input ltp time constants are irrelevant because the dominant pole and its split pole are the only thing that matters in the pass band, and these are virtually identical for a given band width in both vfb and cfb schemes.
rcw
Well RCW the issue is not the first pole from VAS.Is the one from the LTP which is at high f therefore we have then a 90+90 phase shift and a -20-20db/dec ,a second order "filter" . If we have 90 degr and -20db from the vas at the low F (because of the high gain) the ltp will add one pole more at high f which add-90 again thus making the condition for instability when gain is positive .Also shifting in VFB of the pole is good but the phase will suffer .This is regarding the stability .But the phase will suffer anyway .

So a solution with a pole at a higher f or no pole at all is better .The solution can be a common base stage/cascode like CFB has .
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Last edited by catalin; 29th September 2012 at 11:24 PM.
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