| Pierre |
Hello all.
I am building and testing an half-bridge SMPS for +/-55V output. It works very nice so far, although it has been "only" tested up to 125W, but the mosfets and diodes remaing absolutely cool.
It is running unregulated by the moment, and I only have a problem: one of the outputs puts +50V, but the other puts a bit more, around -57V, with the same load.
I can't figure out is causing that assymmetry, as 4 diodes are the same (MUR1560), the inductors (4.7uH) and capacitance (6600uF+100nF) are identical for both rails.
Another thing I would like to do is to measure the switching node voltage with respect to GND. I tried with 4 47k resistors, taking the output from one (1/4 voltage divider), but the RC constant with the capacitance of my 60MHz probe is enough to filter all the details and obtain a rather roundy waveform that doesn't tell me nothing useful. Could I connect the 10x probe directly or is that a bit too much voltage for a typical oscilloscope?
Thanks!!! |
|
|
| Eva |
Are your output inductors coupled? Using two independent output inductor produces poor cross-regulation.
Concerning switching waveform, you can connect your 10x probe to the switching node if it's rated at 400V or more, but be aware that very fast switching transients of high amplitude may cause a bit of partial breakdown in the attenuator that will show as overshoot. I had to live with that problem until I found an affordable 1200V 100:1 probe (there are a lot of good high voltage probes in the market but they are expensive).
You may also build your own high voltage attenuator but you would need to place a high voltaje adjustable capacitor in paralell with the resistors in order to compensate for probe capacitances (that capacitor is the one you can adjust through the calibration screw that most attenuating probes have). |
|
|
| Pierre |
Thanks, Eva.
No, my inductors are independent by the moment.
However, with equal loads and 1 ampere of current flowing, I cannot find the cause that makes the difference in voltages. Can it be an assymmetry in both PWM signals? I don't think so, but... who knows?
Thanks for the advice on the probes also.
Best regards,
Pierre |
|
|
| Eva |
Note that 4.7uH is a very small inductor value, so current ripple will be very high (same applies for voltage ripple).
Also, with such a small inductor value, any slight imbalance in duty cycle between each side of the half bridge will cause a severe output voltage imbalance.
You should try a coupled inductor. If you are using iron-powder toroid cores you can unwind your current inductors, stack the cores, and wind them back together in a bifilar fashion. You would get a dual inductor of approx. 4.7uH (actually 2x 9.4uH when exciting only one winding) by keeping the same turn counts. Additional magnet wire would be required, though.
Theoretically, an ideal dual coupled inductor forces energy transfer between both sides until the voltage across them is the same. |
|
|
| Pierre |
Sorry. The inductors are 47uH, not 4.7uH, They are wound on a drum core, so i will have to substitute them.
Unfortunately, this afternoon I was testing the thing a with a bit more duty-cycle, giving around 140W. Everything was quite cool (the diodes were warm but they have no heatsink), when one of the mosfet exploded.
Prior to this, I had measured the switching waveform by directly connecting my 10X probe. There is no visible overshoot, and the tops are flat. But there is ringing (about 10-20% of the total amplitude) at the dead time. I suppost that's quite normal.
The first thing that has come to my mind is that the same problem that causes the voltage imbalance has caused the failure. The core is a ETD44, with 26 primary turns, the secondaries being 13+13 turns.
Half primary is wound, then the secondaries, then the rest of the primary, to minimize inductance. It was not wound with great care, so perhaps the secondary is not absolutely symmetrical.
Can you point to the possible cause of the failure? Thanks Eva, your help is greatly appreciated!
Pierre |
|
|
| Eva |
Are you adjusting the duty cycle directly with a potentiometer?
Is your prototype mounted in breadboards or something that could lose contact?
Are you sure that your probe didn't slip and made a short?
Have you implemented some kind of current limiting in the primary side? This avoids a lot of catastrophic failures (one of my prototypes that implemented that survived for some time even with one of the switching devices shorted).
Concerning inductors, you may still turn yours into coupled if you wind half of the desired dual coupled inductor on each core in a bifilar fashion and connect them in series. Be careful with polarities. |
|
|
| Pierre |
Yes, by the moment duty cycle is manually adjusted by means of a potentiometer. Maximum setting is about 44%, with 80KHz sw. frequency, 700ns dead-time.
The prototype is built on a double-sided board, with carefull layout, and I think no false contact or short was produced in that moment.
Some details that may give some clue: the mosfets are driven by a IR2110, with 10 ohm gate resistors and antiparallel schottky diode, mosfet are (were) IRFB11N50. The driver has also died.
There is a current sense transformer, whose output voltage is rectified and goes to the shutdown pin of the SG3535, but I don't think that has worked. There is also a 5A fuse before the bridge, but it hasn't blown (one of the mosfet exploded so no shortcircuit was finally produced :xeye: )
My impression is that, as the transformer is coupled by means of a 1uF/250V capacitor, it shouldn't walk into saturation, and the number of turns limits the max flux to an acceptable level, if I have run the numbers correctly.
Thanks for your help! |
|
|
| mzzj |
| Any possibility that you have made 12R and 13R secondaries? |
|
|
| Pierre |
Yes, it is possible.
But with an standard connection of the 4 bridge diodes, would that produce an assymmetry in the voltage of both rails or only an error in the output voltage of both of them with respect with the expected output?
Anyway, I have replaced the driver and mosfets, let's see if it doesn't fail anymore and I can start putting heavier loads. I have discovered that one of the primary ends was not correctly soldered at the transformer pins, perhaps it has suddenly disconnected ¿¿¿???
Best regards,
Pierre |
|
|
| Pierre |
Please verify that the formula I am using for calculation of the flux density vs primary turns is correct:
Np=(V*10^8)/(4*B*f*A)
where...
V=320/2=160V
B is expressed in Gauss
f is expressed in Hz
A is ¿EFFECTIVE AREA? in cm^2 (172mm^2=1.72cm^2 for ETD44) ???
That gives 1100 gauss for 26 primary turns, at 80KHz that's about ok for a 3C90 core, isn't it?
Thanks for the clarifications! |
|
|
| Eva |
Asymmetric secondaries won't produce asymmetric output voltages in that configuration since each secondary is powering both output rails alternatively.
Output imbalance is probably due to a duty cycle asymmetry issue that the DC blocking capacitor compensates with some offset, thus yielding asymmetric volt*second product to the output inductors.
On the other hand, I've been using SG3525A for several years and some units have proved to be quite asymmetric, particularly when a low value discharge resistor is employed and at low duty cycles. This behaviour suggests that the IC is actually suffering from internal ground loops, but that shouldn't be an issue when coupled output inductors are employed.
Note that the asumption that duty cycle will be perfectly matched between sides of the bridge can't just be made because there are several unpredictable sources of mismatching and the circuit must cope with them. |
|
|
| Pierre |
Thanks, Eva. That matches with what I thought: differences in secondary turns cannot produce different in output voltages in each rail.
I hope that won't cause major problems like transformer walking into saturation (you have worried me saying that the coupling cap can create offset due to different duty cycles)
I will measure the duty-cycle of both SG3525 outputs carefully to see if there are any differences for various duty-cycles. If I find them, I will try with different discharge res. values to see if that's corrected.
Your experience is unvaluable!
Best regards,
Pierre |
|
|
| Pierre |
Done: the SG3525 outputs (with the mains disconnected and no mosfets, that shouldn't change anything, anyway), have duty-cycles that differ only in a decimal, for example, 30 and 30.1%.
The greatest difference is at max. duty cycle: 44% and 44.3%. That kind of error may be due to my Tektronix DSO, however. In any case, that should produce a difference of less than 1V, while I have observed up to 7-8V unbalance.
Now I have to check the gate-source waveforms of both mosfets just in case there is something weird in the high side or whatever.
so... having an assymmetric secondary (12 vs 13 turns, for example), shouldn't have any effect other than output voltages different to calculated (but both the same), right? I was thinking in re-winding my transformer, but if that's true, I think it doesn't worth the pain.
Thanks |
|
|
| mzzj |
| quote: | Originally posted by Pierre
Please verify that the formula I am using for calculation of the flux density vs primary turns is correct:
Np=(V*10^8)/(4*B*f*A)
where...
V=320/2=160V
B is expressed in Gauss
f is expressed in Hz
A is ¿EFFECTIVE AREA? in cm^2 (172mm^2=1.72cm^2 for ETD44) ???
That gives 1100 gauss for 26 primary turns, at 80KHz that's about ok for a 3C90 core, isn't it?
Thanks for the clarifications! |
flux swing (delta)B=(V*t)/(N*A)
B=Teslas
V=160v
t= maximum pulse width in us
N=turns
A=efective core area in mm^2
I got flux swing of 220mT wich is same as your 1100 gauss peak. i prefer this equation as it is easier to remember for me and uses easy units without coefficients :)
110mT on 3C90 @80khz should be ok. |
|
|
| Eva |
| The DC blocking capacitor charges in order to compensate for asymmetric duty cycles and prevent saturation, but that correction offset, together with the asymmetric duty cycle that it's compensating for, are a source of output imbalance. The DC blocking capacitor will prevent any kind of saturation as long as fixed duty cycle or voltage control is employed (it will cause a lot of trouble with current control, though). |
|
|
| Pierre |
This is starting to resemble a chat! ;-)
Well, then it seems that everything is about right, and that mounting a coupled inductor and paying attention to the duty cycle difference will solve these problems.
I will try to post some photos of the primary switching waveform for you to see it: I am only a bit worried about what's happening the portion of time that both mosfets are off, where you can see some damped ringing (I have to measure the freq. but it is several MHz for sure), with around 80Vp-p maximum amplitude. I don't know if that's dangerous or causes dissipation, or if it should be corrected before going to higher power testing.
Best regards |
|
|
| Pierre |
Unfortunately, my mosfets have exploded again, as soon as I have increased output power from aroun 100W to about 170W. Now I had 2sk2141, 6A/500V, more than enough for that power.
I attach a figure of the switching waveform when it worked (100W output, worked fine and everything cool), with around 20% duty cycle. It is something strange, isn't it? Y scale is 50V/div, X scale is 2us/div
What can be the cause of the violent explosion (they weren't hot, in fact they exploded just when I plugged the supply)? I am quite puzzled. |
|
|
| carvinguy |
This has been a very interesting discussion so far! (Sorry, Pierre, for all the smoke...)
I would perhaps suspect the MOSFET drive circuitry and check that out once again. The IR2110 type driver parts are very picky to set up correctly. I've tried to use them before and ended up going with a transformer drive which is much easier to deal with (to me, at least). If the IR part is not bypassed REALLY well and everything with nice tight loops, you may be getting ringing above or below the Vgs limit of the FETs(usually around +/-30V). The layout problems (if there are any) and ringing will increase when you start to increase the load.
If your FET's blew at turn on, maybe you had FET cross-conduction because of the driver chip not establishing a stable operating point yet. If I recall correctly, though, the chip has UVLO, so would not have turned on without an adequate voltage. Strange indeed...
Something to think about...
I'm curious about the ringing in your picture, too, and hope someone (Eva ;) ) can shed some light on that.
Matt. |
|
|
| Eva |
The waveform looks right. It shows discontinuous mode operation. Output inductors are charged during the on time, then they discharge during the part in wich voltage rests at center (inductors are effectively shorting the secondaries), and then the transformer is left free so it tends to reset from the previous cycle but the discharged inductors in the secondary side clamp the voltage.
As carvinguy said, this may be an issue with gate drive and resonance peaks exceeding maximum Vgs rating, thus punctuting metal oxide gate isolation and inmediately destroying the device (and the gate driver IC).
It may be also an issue with continuous conduction mode and secondary side diode reverse recovery, since according to the 100W waveform 170W semms like a bit past the continuous mode boundary. Note that if there is a gate drive issue, it may arise only in these circumstances.
You will need to connect oscilloscope probe directly at MOSFET terminals and build an improvised common-mode filter for the probe like the ones shown in the picture in order to get an accurate picture of gate drive waveforms.
 |
|
|
| Danko |
Can it possible, that the high-side floating supply (bootstrap-capacitor) can't charge fast enough, and the low side FET is already switching, while the upper side mosfet does nithing, becouse the bootstrap-capacitor isn;t charged up properly, and the IR2110's UVLO protection is disabling the ouput?
ups, nice sentence :) |
|
|
| Pierre |
It is true that IR2110 are tricky to use. I have a long experience using them in Class-D amplifiers (have a look at "Mosfet reliability in Class-D amplifiers" thread, or something like that, under "Class-D" forum).
First, a lot of mosfets blew up instantaneously just like in this supply, with no apparent reason. I rearranged the layout paying great attention to the gate loop and used more rugged mosfets (moved from OnSemi to IRF) and since them I have had no problems, and I have tested them quite a bit!
All that experience has been applied here, so my layout, at least in that respects to the IR2110 chip and gates should be ok. Anyway, I was cautious enough to add a double footprint in order to add a gate drive transformer in the same board, that I will test soon (I am fed up with so much mosfet and drivers changes!)
This way even the SG3525 chip seems to have died.
If the problem was excessive Vgs voltage due to resonance, what about adding 15V zeners in parallel with Vgs? I know they are not recommended for high frequency, but at 80 KHz...
But if it is due to spurious turn on causing shoot-through, that's a different story...
BTW: Danko, my bootstrap capacitor is 10uF, should be enough. Gate resistors are now 15 ohm. |
|
|
| Pierre |
I have changed the IR2110 to a SD250-4 Coilcraft gate-drive transformer. I still haven't soldered my mosfets, but with 1nF capacitor load, the Vgs waveform looks clean (bipolar, +/-15V amplitude, and a rise time from 0 to 15V of around 350nS, I assume that this will produce much faster rise time in the mosfets, as the Vth level is reached in around 50-80ns)
My mosfets are rated at +/-30v Vgs voltage, so they shouldn't be damaged by the +/-15V bipolar wavefrom... I will cross my fingers.
This afternoon I will test all and tell you how it goes. At least I won't have to change my chips once a day during my testing phase.
Pierre. |
|
|
| Pierre |
More about my progress....
I exchanged the IR2110 by a GDT (Coilcraft sd250-4). The figure shows the waveform of Vgs. It is clean although a bit slow for my taste, and there are a peak that may trigger the mosfet (although the other will be off so I don't think that's an issue).
I have been able to obtain 375W with this and observe a good switching waveform. I haven't tried more but I am sure it will do well. I am almost sure that the mosfets heat up more quickly than with the IR2110 driver, I suppose that the rise/fall times are slower, so
I will try to add a NPN/PNP buffer for the SG3525 to speed up things a bit.
I have finally decreased sw. freq. to 40KHz, and now they become less hot, so switching losses must be dominating.
Unfortunately, when I have soldered the transformer and everything ok, I have done an stupid mistake and something has exploded and broken (but not the mosfets) But as soon as I have repaired all, I will try to add regulation. |
|
|
| Pierre |
| And this is the switching waveform with 200W load at 80KHz. It still shows discontinuous mode. |
|
|
| Danko |
| How would the waveform look like in continous mode? |
|
|
| carvinguy |
Yes, the SG3525 doesn't have a whole lot of drive current available, so rise times are slower than what you'd like because it takes longer to get charge to the gates. The IR part has 2 amps available for each device compared to 400mA for the SG part. I think your results will be much better after adding the additional driver circuitry.
Matt. |
|
|
| Eva |
The transformer gate drive waveform shows that the transformer is saturating and that leakage inductance is already limiting MOSFET turn-off speed, so it can't be improved. There may be also some cross conduction due to the leakage inductance spike.
You can't just drive MOSFET gates directly from a transformer and expect low losses, this only works with IGBTs that have input capacitances almost an order of magnitude lower than MOSFETs, and with bipolar transistors that are current driven.
It would be a good idea to add some kind of gate buffering, at least for turn-off. Don't try to load the transformer with a lower gate resistor to get faster turn-off since this will only make the leakage inductance spike issue worse. |
|
|
| Pierre |
Eva, you are right in that last affirmation: I already tried to lower the gate resistor. Now is 15 ohm but I tried up to 0 ohm, and yes, it switches a bit faster but the spike rises, so I didn't try with the mains switched on for security.
So, from what you are saying, if I keep the mosfets, I should look for another transformer, right? Can you suggest any commercial model that could work, having a look at the specifications of SD250-4?
Thanks! |
|
|
| Pierre |
mmmm.
Have been thinking about the GDT a little bit more. Eva, how do you know that it is saturated?
It is rated at 375V-us, so as I drive it in a bipolar fashion, (+/-12V), at 40 or even 80 KHz, there shouldn't be any problem, right?
If it is still usable at that frequency, how can I speed it up to minimize losses?
Another question: can I simply substitute the mosfets by IGBTs (properly rated and fast enough for that frequency, of course)?
Thanks! |
|
|
| mzzj |
| quote: | Originally posted by Pierre
mmmm.
Have been thinking about the GDT a little bit more. Eva, how do you know that it is saturated?
It is rated at 375V-us, so as I drive it in a bipolar fashion, (+/-12V), at 40 or even 80 KHz, there shouldn't be any problem, right?
If it is still usable at that frequency, how can I speed it up to minimize losses?
Another question: can I simply substitute the mosfets by IGBTs (properly rated and fast enough for that frequency, of course)?
Thanks! |
From drooping dc level on your gate pulses. 10 different choices also why it is happening, I dont know your exact part values and scematics so difficult to say. Do you use dc-decoupling cap on GDT primary? too small dc-decoupling cap or too small chip vcc cap can cause similar dc pulse drooping as core saturation.
I dont see any tricks to speed it up, lower leakage tranny is only option if your layout is already reasonablyy good. I have had good results with DIY gate drive toroids, sub 50ns risetimes to 6nF load with minimal ringing. Sorry, cant recommend any off-the-self products. |
|
|
| Pierre |
The thing is that, the more I look at the specifications of the SD250-4 transformer I am using, the more convinced I am that it is perfectly suitable and that it should produce much better results.
It has relatively low primary inductance: 1.5mH and leakage inductance (4 uH max), 375 V-us ET product (fast enough), specified for 10-250KHz use, I am at 40 or 80 KHz... Why should it be saturating?
I am driving it in a bipolar fashion directly from the SG2525 chip, with a 4.7 ohm resistor in series with the primary (no coupling cap), and 15 ohm resistor in series with each gate.
And the mosfets are not specially slow: 52nC max gate charge, 1400pF max input capacitance.
The tracks are quite short. Am I doing something wrong? |
|
|
| Eva |
MOSFET turn-off is a delicate process because a 300V positive-going pulse is coupled to the gate through reverse transfer capacitance. That's why driving MOSFETs from transformers produces high losses and sometimes cross-conduction.
The usual approach is to use a PNP emitter follower and some additional components to buffer turn-off process. Leakage inductance is quite hard to reduce, and it increases with volts*seconds product rating.
Also, IGBTs will be a direct replacement for MOSFETs and they will actually switch faster with the same transformer and gate resistors because their capacitances are an order of magnitude lower.
You may also try to solve the gate drive IC issue. If it's a gate overvoltage issue, Zeners and schottkys soldered directly across gate and source terminals may solve the problem.
Could you show us your schematic and layout? There may be some non obvious mistake that you are overlooking. |
|
|
| Pierre |
Hello, Eva. Nice to hear from you again.
Here is my drive stage schematics. Forget the NPN/PNP buffer because it is not implemented by the moment, although I would like to try it at some moment. I have ordered a couple of samples of Pulse GDT's, let's see if they work better.
I have selected this IGBT, do you think it can be suitable for 40 to 80KHz operation? : IRG4BC20W |
|
|
| Pierre |
Now I have another problem: I am trying to implement regulation, using an optocoupler (see attached schematics).
The problem is that, while adjusting the potentiometer, R16, (that does vary the "ERROR" node voltage from 0 to around 9V), it cannot make the PWM go lower than maximum duty-cycle, that is, the output of the error amplifier inside the SG3525 chip doesn't go lower than about 3.7v (max. duty cycle corresponds to about 3.5v).
In fact, I have tested with no power, simulating the NFB loop by putting a variable voltage at the "ERROR" node, and as I increase it, the error amp. output goes down to about 3.7V and there is a point where it starts going up again, so I can never reach PWM less than around 50%. Strange, isn't it?
I think it is an issue with the reference applied to NI input or something like that.
Best regards,
Sergio |
|
|
| Eva |
IGBTs don't have a body diode like MOSFETs so you should either solder a diode between collector and emitter or get an IGBT with built-in diode.
IRG4BC20W doesn't include a clamping diode but otherwise it would be fine. Note that reverse transfer capacitance is less than 10pF. Also note that IGBT conduction losses at low currents will be higher than in MOSFET, but they won't skyrocket as current is increased like in MOSFETs. |
|
|
| Pierre |
You are right... unfortunately, IGBTs with diode are difficult to find and much more expensive. I will give mosfets a further try.
When I have solved the regulation issue, I will continue with this.
Best regards,
Pierre |
|
|
| Eva |
| I bought a bunch of SKP10N60 from RS-Amidata. They have a built-in diode and I don't think they were expensive. |
|
|
| Pierre |
Yes, Eva, they seem good enough and not very expensive. Thanks.
About regulation, I have built a small prototype board with the sg3525 chip just for testing, and found that the problem was that the inverting input gain (resistor between pins 1 and 9) was too low: 1.5 V/V (R was 15k and input R was 10k), so the error opamp output (pin 9) was doing strange things (as not inverting for some input ranges).
Once I have increased that resistor to >150K (G>15), the circuit responds linearly with increasing duty cycle for decreasing voltage and vice-versa. All this in a very limited range of about 300mV of input voltage around 5.1v (Vref): Below around 4.95V, PWM is 47%, and above around 5.25V, PWM is 0%.
I will try to make the modification in my SMPS and test if it regulates correctly.
Best regards,
Pierre |
|
|
| Eva |
SG3525A suffers from error amplifier latching when inputs are allowed to go below 1V above groud or so. I stumbled into similar problem because I'm using this error amplifier for average current control, and I solved it by adding a 2.5V bias to both signals.
There are other low-cost ICs like TL494 whose common mode input range includes ground, but TL494 has its own drawbacks in comparison with SG3525, particularly it lacks double pulse supression and premature pulse termination capabilities.
Also, be very careful when loading COMP output of SG3525A, it has a very low sink current capability, something like 100uA, so load impedances below 50K may be troublesome. Try to use that amplifier as an unity gain buffer and place the actual error amplifier and its compensation in the secondary side instead, before the optocoupler. |
|
|
| Pierre |
Ok, thanks for the suggestions.
As a first approach, I will use the SG3525 error amplifier in the primary side and only zeners+resistors in the secondary, as shown in the schematics I posted. Let's see how it works then. Once it is up and running, I will try to improve it.
As for the coupled inductor to balance both outputs, I am planning to stack two T106-2 cores (micrometals) and bifilar wind about 40 turns for 20uH, with 1mm wire. Then I will use one wire for each rail, but in opposite directions.
Hope this work well for the kind of power I am trying to achieve.
Best regards,
Pierre |
|
|
| Pierre |
I have been, at last, able to regulate my PSU.
The control setup is as follows:
- Secondary: two 30V zeners, 1k5 resistor and optocoupler diode in series, from + rail to - rail.
- Primary control: optocoupler phototransistor has collector to +12V (control IC supply), emitter with adj. resistor to GND (filtered with a 120nF cap as I found it very noisy)
- Internal SG3525 Error opamp configuration: Error signal is taken from phototransistor emitter, fed to the error opamp inverting input via a 10K resistor. From there, 330K resistor in parallel with 10nF, to COMP pin (inverting amplifier with gain 33).
The non-inverting input is tied to Vref (5.1V)
COMP pin has about 500nF to GND.
The supply regulates OK, but with "light" loads (light being 100W), it shows some strange periodic oscillation, about 30Hz, 4Vpp (amplitude and freq. varying as load changes).
It is as it the control was not very estable. I am also afraid that with those values its transient response will be veeery slow.
How can I find the optimum values for both (almost)no load good behaviour and adequate transient response?
The supply is running at 40KHz by the moment. Max power I have tested is 450W for several seconds.
Best regards,
Pierre |
|
|
| e96mlo |
Won't you get the tolerances of the zeners on the "wrong side" of the regulation this way?
Your error amp will never know if it is regulating a zener missbehaviour or the correct output voltage. |
|
|
| Pierre |
Well, of course, from unit to unit there would be an error due to zener tolerances, but once it is adjusted (there is a potentiometer), the output should be reasonably estable (without having into account temperature effects on the zeners, optocoupler and references, of course).
But that should be more than enough, anyway.
The problem is that it tends to "oscillate", and I have found that not only at light loads. What changes with load is the frequency of that oscillation. Its amplitude is around 4Vpp.
Best regards |
|
|
| Eva |
| Why don't you wire the error amp in the way I told you? That 120nF capacitor is ruining stability, there must be only a single LF pole provided by the output filter and compensated by a zero in the error amplifier, and the optocoupler is probably too slow to keep the required phase margin at high frequencies when compensation is not applied in the photodiode side. |
|
|
| Pierre |
Thanks for the comments.
Sorry Eva, what was your suggestion: using an external opamp and keep the SG3525 internal error amplifier as a non-inverting unity gain buffer?
I added the 120nF capacitor at the start, in open loop (when I regulated the output by means of a potentiometer), because when I observed the emitter waveform it was pulsing at the switching frequency, when I expected DC (with noise) with estable dummy load and input voltage, so I added it as a lowpass filter.
Now it comes to my mind that perhaps it was due to noise pickup by the scope probes...
I will try to remove it and see what happens.
Given that my LC filter is 47uH + 6600uF per rail, what freq. should I place a zero in my fb loop at?
Thanks! |
|
|
| Pierre |
After a little more experimenting, I have decided to follow Eva's advice and add an external opamp for the error amplifier.
My PSU regulates about ok, but there is a low frequency ripple (around 5-6Vpp) with low loads, which produces a very audible sound on the amplifier. I guess it is not estable.
Please tell me if the following sch is a good starting point. The error amp must be compensated, but first let's concentrate on the DC configuration:
thanks!!! |
|
|
|
