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Old 27th September 2005, 09:11 AM   #31
mzzj is offline mzzj  Finland
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Quote:
Originally posted by Eva
As opposed to low voltage ones, high voltage MOSFETs are quite expensive those days and prices don't seem to fall... rather they rise

Also, paralelling is almost always required for high power applications, thus multiplying the cost. I don't understand why engineers insist in using them everywhere, even in those designs that doesn't require extremely fast switching (like mine).
Going to be SERIOUS power supply if someone needs to parallei IPW60R045's

Yeah, mosfet prices seems getting higher, but look what you get compared to old-fashioned mosfets...
IRFP460 vs. IPW60R045
Qg 210nC vs. 190nC
rdson 270mohm vs. 45mohm!

Similar(little better) gate charge but one-sixth on-resistance.

Or more reasonable comparision between IPP60R199CP similar rds-on competitor, 190 mohms and 43nC
Might be worth of couple of extra $$ as gate drivers can be minuscule compared to IRFP460

----
I guess engineers are scared of relatively complex lookalike base drive and bigger assembly/parts cost. On the other hand almost all AT/ATX power supplies are with NPN-bipolars, so they still have their places....
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Old 28th September 2005, 05:56 PM   #32
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Quote:
Originally posted by Eva
As opposed to low voltage ones, high voltage MOSFETs are quite expensive those days and prices don't seem to fall... rather they rise

Also, paralelling is almost always required for high power applications, thus multiplying the cost. I don't understand why engineers insist in using them everywhere, even in those designs that doesn't require extremely fast switching (like mine).
I guess there are several reasons for the
preferred use of the MosFets, also in slower applications:
1.) You don't need continuosly high drive currents.==> Less system complexity, because proper base drive without transformer is often
difficult.... (and base drive with transformer is always difficult for less freaky people... )
2.) Avoiding trouble with different storage times of different production lots.
3.) Integrated body diode helps to reduce the system complexity.
4.) Usually quite comfortable SOA, while with BJTs you have to look about 2ndary breakdown. Also max. voltage rating depending on the base drive.
5.) Avoiding trouble with reduced current gain at high currents.

But I agree that BJTs are still often a good choice. Not only regarding price. Also the losses at higher currents are quite attractive.
And in fact they are often used in switching applications!
ATX PSUs
Lighting (sometimes)
Motor Drives (OK, IGBT...)
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Old 28th September 2005, 07:22 PM   #33
Eva is offline Eva  Spain
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With these bipolar transistors, SOA and RBSOA are no longer a concern, check these figures:

Click the image to open in full size.

Forward bias SOA is almost square for pulses up to 10uS, so it's quite hard to exceed, not only during turn-on, but even during transient overload conditions where not enough saturation is achieved. Actually, the current limiting feature of the older 1KW voltage-mode control circuit wasn't working properly and it was allowing for 25A transients without bipolar transistors failing.

Also, reverse bias SOA only applies during crossover and fall periods, when the base junction is reverse biased and the device is still conducting. However, proper base drive and RC snubbering of the inductive load actually allow to safely ignore RBSOA:

First: The 33 ohm and 2nF snubber (actually 25 ohm) across the transformer increases full-load crossover time from approx 40ns to 100ns, so with a 250V supply it is still capable of sinking all the current from transformer leakage inductance. And the higher the supply voltage the more current it can sink, thus ensuring that bipolar transistors stop conducting far before the voltage reaches the opposite rail, for I'm considering a PFC front-end with 400V rails to ultimately extend output current to 180A or 200A.

Second: The leakage inductance of the base drive transformer provides a nice -12V peak (actually clamped by reverse breakdown of the base-emitter junction) during the entire crossover period and even longer. Actually, the maximum posible reverse bias voltage is guaranteed for loads above 50% (Ic=5A and up).

This means that turn-off is safe even for Ic=20A, while average current limiting prevents it from switching more than 11A average, and an auxiliary peak current comparator with slope compensation prevents peak currents over 14A during fault conditions. It has even survived shorted transformer secondaries for several seconds from time to time.

As opposed to all those common myths about bipolar transistors, my only concern about them is currently excessive turn-on losses due to slow saturarion and inherent limitations in the transformer-coupled base drive scheme.

I will post some captures from the oscilloscope soon...
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Old 28th September 2005, 08:48 PM   #34
Eva is offline Eva  Spain
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These are the captures:

Click the image to open in full size.
Switching waveforms at idle, red trace is collector voltage at 50V/div, blue trace is measured current at 2A/div, timebase is 5us/div.

Click the image to open in full size.
Switching waveforms with approx. 60A output, red trace is collector voltage at 50V/div, blue trace is measured current at 2A/div, timebase is 5us/div. The bus voltage drops badly due to my isolation transformers. Note the big spike due to the RC snubber.

Click the image to open in full size.
Switching waveforms with approx. 125A output, red trace is collector voltage at 50V/div, blue trace is measured current at 2A/div, timebase is 5us/div. It was captured very quickly after the load was switched in order to avoid showing extreme voltage drop.

Click the image to open in full size.
Detail of turn-off waveforms with approx. 115A output, red trace is collector voltage at 50V/div, blue trace is measured current at 2A/div, timebase is 200ns/div. This one shows the huge bus voltage drop due to loading 1KVA transformers with 1,8KW of rectified load. Note that the apparent current tail is not due to the bipolar devices at all, actually they stop conducting quite quickly, instead the tail is just a measurement artifact caused by the magnetic snubbers being trapped between two diodes (it disappears if the inner clamping diode diode is removed). This capture also shows losses due to slow saturation, an issue that comes more from the limitations of the single transformer base drive approach than from the nature of the bipolar devices themselves.
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Old 28th September 2005, 09:05 PM   #35
Eva is offline Eva  Spain
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Click the image to open in full size.
Base waveforms with approx. 125A output, red trace is base-emitter voltage at 2V/div, blue trace is measured current at 2A/div, timebase is 5us/div. Note the negative peak at turn-off and the clamped peak at turn-on that is limiting the base current rise slope and slowing down saturation (clamping comes from the pair of MOSFETs with Vds=55V maximum that short the base drive transformer, whose turn ratio is 7:1, altough I'm considering leaving the poor BE junctions clamping those peaks alone).

Click the image to open in full size.
Detail of base waveforms with approx. 125A output, red trace is base-emitter voltage at 2V/div, blue trace is measured current at 2A/div, timebase is 200ns/div. Note the 12V negative base voltage peak that ensures the best RBSOA possible.

Click the image to open in full size.
Base waveforms with approx. 125A output, red trace is base-drive cell current (including the damping RC network) at 1.5A/div, blue trace is measured current at 2A/div, timebase is 5us/div. Note the negative Ic/2 peak with controlled slope at turn-off, and also the positive peak generated by my "saturation-boost" circuit at the beginning of the cycle, that does not rise fast enough in order to reduce the "on resistance" of the bipolar transistors quickly enough.
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Old 28th September 2005, 09:17 PM   #36
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Just for my own clarification:
Your blue traces are showing the current of your primary transformer...
....measured and rectified as seen at I_sense in the schematic ?
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Old 28th September 2005, 09:29 PM   #37
Eva is offline Eva  Spain
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Blue trace is the current measured through the magnetic snubbers, it's rectified due to the topology of the measurement system and it includes the current from the primary winding of the main transformers, the current from the RC snubber and the clamping tail from the inner emitter/collector diode. The trace itself is the voltage drop across the 100 ohm resistor that is connected to the collectors of the cascode transistors.

Multiply that value from the blue trace by 12 and you will get secondary winding current, and output inductor current will have almost the same value, except by the peak at the beginning of the cycle caused both by RC snubbers and by the slight clumsyness of low-cost MOSFETs used as synchronous rectifiers (those actually suffer from tail currents!!)

Actually, the voltage from this blue trace is fed directly to the auxiliary peak current comparator, and it's also further processed in order to simulate the inductor current downslope and finally it's fed to the average current amplifier that controls duty cycle and to the "synchronous rectification enable" comparator.
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Old 28th September 2005, 10:23 PM   #38
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...I really love this base drive.

But I am not sure if I would dare to implement your proposal
of a 400V PFC rail and 180A output at the same time.
Directly after BJT turn OFF, when Uce wheels up (worst case over shoot up to full rail) the Ube is just kept around zero by a resistor.
The negative base drive peak is available until Ic has sloped down, but not longer. Your pictures do not exactly show Ube and Uce, but I would guess that max. Uce overshoot takes place about 100ns-200ns after Ic has sloped down. Then during this overshoot probably no negative base drive is available anymore.
At higher rail voltages the BJT might need some ongoing neg base drive until Uce overshoot has calmed down in order to overcome
internal BJT leakage (above 350V) and capacitive impact from Uce sloping to Ube.
...may be I am a coward...
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Old 28th September 2005, 10:35 PM   #39
Eva is offline Eva  Spain
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I explained it previously. There are 1.2V of permanent negative base bias during dead time, plus up to 2 additional volts for Ic=10A. This means that after the -12V peak the waveform rests at approx -3.2V, not 0V.

Also remember that the collector voltage rise is heavily slowed down by the RC snubber, the transistors stop conducting at all when the collector voltage is still half way to the opposite rail.

And note that for RBSOA it's collector current what matters. After Ic has dropped to zero, the device can sustain 700V of Vce as long as its base is kept at the the same or a lower potential than the emitter.
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Old 28th September 2005, 10:43 PM   #40
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Quote:
Originally posted by Eva
This is the base-drive cell:
Click the image to open in full size.

When the switching transistor is turned on, the drive transformer provides as a base current 1/5th of whatever collector current is flowing. That 10uF capacitor is charged to two diode drops (1.2V), then the PNP transistor starts conducting and the 10uF capacitor charges further depending on the voltage drop that the base drive current causes across the 1ohm resistor (2V for Ic=10A).

...got it. Finally, got it.
Great !!
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