0-15V 0-120A adjustable PSU attempt with average current control. - Page 3 - diyAudio
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Old 12th September 2005, 05:13 PM   #21
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Eva, could you please contact me off-line directly, regarading a power supply project, at:

hans14914 (at) yahoo (dot) com

Many Thanks
Hans
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Old 24th September 2005, 05:20 PM   #22
Eva is offline Eva  Spain
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I have designed and built a new full bridge PCB. This is how it looks:

Final layout:
Click the image to open in full size.

Partial unfinished layout (placing the storage capacitors on the other side produced better matched track lenghts between each phase of the full bridge, thus providing more symmetrical drive to the transformer):
Click the image to open in full size.

Final layout in detail. Q5, Q6, Q7 and Q8 are not switching devices, they are auxiliar 60V 4A PNP transistors used by the base drive circuit. I will show the schematic of the "base drive cell" later:
Click the image to open in full size.

Etched PCB (I had to use two joint pieces since I ran out PCB material):
Click the image to open in full size.
Click the image to open in full size.

Detail of the magnetic snubbers that I use both to get a precise measurement of collector current of each transistor and to prevent reverse biasing and parasitistic turn-on of them. Initially I used 1:7 turns but that was not enough to prevent core saturation, so I increased it to 3:21 turns and it worked fine:
Click the image to open in full size.
Click the image to open in full size.

Detail of the 10uF 10V SMD ceramic capacitors that I use for base drive, they are tiny yet they feature very low ESR and no ripple current constraints. They are subject to repetitive peak currents in excess of 6A and don't complain:
Click the image to open in full size.

Overall view with most components mounted. The base drive transformer is intentionally shifted to the left in order to make the PCB tracks of the right side longer and increase their resistance, with the purpose of compensating an inherent transformer winding asymmetry issue that forces the magnet wires of the right side to be a bit longer (Currently I have replaced that EF20 transformer by a bigger ETD29 since magnetizing inductance was too low. The asymmetry issue remains but requires far less compensation. I will explain in detail later):
Click the image to open in full size.

Detail of mains diode bridge, that was quite an improvisation since I hadn't any suitable heatsink and I had very little available PCB space:
Click the image to open in full size.

Overall view with all components mounted and ready for testing:
Click the image to open in full size.

Another overall view, note how bulky the mains rectification section has become in order to power a 2KW load with reasonable ripple in the +300V bus. The EMI filter is also shown, a common mode choke and a 470nF capacitor appear to be more than enugh to prevent any line interference:
Click the image to open in full size.

Another more overall view. I have placed the resistors from the RC turn-off snubbers in the "wind tunnel" side so that more than 2W could be dissipated in conventional low-cost 2W units without overheating.
Click the image to open in full size.
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Old 24th September 2005, 07:04 PM   #23
Eva is offline Eva  Spain
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This is the base-drive cell:
Click the image to open in full size.

When the switching transistor is turned on, the drive transformer provides as a base current 1/5th of whatever collector current is flowing. That 10uF capacitor is charged to two diode drops (1.2V), then the PNP transistor starts conducting and the 10uF capacitor charges further depending on the voltage drop that the base drive current causes across the 1ohm resistor (2V for Ic=10A).

At the end of the cycle, the drive transformer is shorted and base current progressively changes from being sourced into being sunk at a slope controlled by the leakage inductance of the transformer. That slope has a optimum value that produces the smallest collector tail current and the most abrupt increase in Vce, yielding the lowest losses. The peak negative base current is not only limited by leakage inductance, but also by the 0.3ohm resistor (three 1ohm in paralell) and the voltage across the 10uF capacitor, so that it has a minimum value that increases for increasing collector current levels.

When the switching device is turned off, the PNP transistor prevents the 10uF capacitor from discharging through the 1ohm resistor. This means that duty cycle doesn't affect the amount of negative base current available at turn-off, it only depends on actual collector current wich is highly desirable. For converters operating at almost fixed duty cycles this PNP transistor may be safely replaced by one or two 1N4001 diodes (in fact, it's not included in AT nor ATX power supplies), however, I want the output to be reliably adjustable.

The 1K resistor prevents parasitistic turn-on due to the inherently high leakage currents present in high voltage bipolar transistors, and the 47 ohm resistor speeds up the turn-off of the PNP auxililiary transistor. Finally, the RC network provides some damping for ringing and spikes.



This piece of schematic illustrates the magnetic snubber scheme in use (NOTE: These transformers are shown with the wrong polarity, one of the windings should have the point in the opposite terminal. Sorry):
Click the image to open in full size.

Each transformer prevents reverse biasing of the corresponding bipolar switch, wich is required because clamping diodes are not fast enough to prevent some degree of reverse biasing and reverse turn-on of the switching transistor (remember that bipolar devices also work with E and C pins exchanged, but in a clumsy and very slow way, causing a long current tail after the spike). Additional clamping diodes are included in order to protect the bipolar transistors from leakage inductance spikes produced by the own snubbers.

The 1:7 ratio was chosen because it allows to use a small 1A diode in the secondary side while still keeping to a minimum extent the diode capacitance multiplicacion effect that the transformer provides. With too much reflected capacitance the snubber would show a low impedance and would fail to prevent reverse biasing.

That sample of 1/7th of collector current is further employed to measure output current as that schematic shows (these ones have the right polarity):
Click the image to open in full size.

The 1/7th of Ic sample is summed with the sample from the partner transistor by means of an additional 1:1:28 current transformer, thus obtaining a 1/100th (actually 1/98th) sample of main transformer primary current. Cascode transistors are employed in the secondary side in order to keep voltage across the transformer below 1V for all current levels, thus allowing to use very small ferrite cores while still obtaining a gentle 1V/1A noise-free signal.

(More to come).
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Old 25th September 2005, 01:34 AM   #24
Mark Kravchenko --- www.kravchenko-audio.com
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Thumbs up Coooool

Looking good EVA!

MArk
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Old 25th September 2005, 01:10 PM   #25
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... not so bad your tiny toy..
OK... OK.. AMAZING

Quote:
Originally posted by Eva

At the end of the cycle, the drive transformer is shorted and base current progressively changes from being sourced into being sunk at a slope controlled by the leakage inductance of the transformer.
... so you are shortening the transformer not on the secondary.
Just curious about details how you feed the control signal into the base drive cell...
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Old 25th September 2005, 04:49 PM   #26
Eva is offline Eva  Spain
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No, the transformer is shorted in its secondary winding by means of two MOSFETs.

I'be veen considering a new system for some time, where these MOSFETs would be placed directly in the base drive cell, but it requires a pulse transformer for gate drive and another transformer to provide gate power to all the cells, so that the system can stay turned off. Also, a third pulse transformer would be required to drive four additional MOSFETs for my "saturation boost" system (I will explain it later).

On the other hand, I've solved most of the issues that I had with fully transformer-coupled base drive, except the inherently limited base-current rise slope at the begginning of the cycle, that slows down saturation (transformer voltage is clamped in the secondary side in order to protect the reverse biased B-E junctions from excessive breakdown current). However, reducing leakage inductance by adding more layers to the transformer increases turn-off losses.

Finally, I have the full-MOSFET alternative: SPP20N60S5 from Infineon. With eight units of these (four groups of two) I would get the same conduction losses as with bipolars, with slightly lower turn-on losses and similar turn-off losses. However, I would have to drive eight 4nF gates, thus requiring something like eight 2A gate buffers, and each of these devices would cost me almost as much as 4 bipolar transistors. (And in these circumstances IGBTs have little or no advantages over bipolar transistors).
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Old 25th September 2005, 05:24 PM   #27
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If you shorten the secondary: How does the leakage inductance
affect the negative base current?
Two Mosfets?
...do I see a center tapped secondary?

SPP20N60S5:
a) ...you found it
b) This one might match even better :IPP60R099CS
http://www.infineon.com/upload/Docum...9CS_rev1.1.pdf
c)Yes, your bipolar design seems to be perfectly competitive in technical terms. Competitive vs. top rated expensive MosFets!
And as your are not going to produce millions of SMPS, you don't need to fear the tolerances of the BJT storage time or any other components.
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Old 26th September 2005, 08:46 AM   #28
mzzj is offline mzzj  Finland
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Quote:
Originally posted by ChocoHolic

SPP20N60S5:
a) ...you found it
b) This one might match even better :IPP60R099CS
http://www.infineon.com/upload/Docum...9CS_rev1.1.pdf
Do you have distributor for those if I want to buy less than 500pcs? IPW60R045 seems also intresting but samesame problem, i can find only 240 pcs. min order and 14 euros apiece

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Old 26th September 2005, 09:05 PM   #29
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I am sorry. I am not with Infineon, also no 'side chanels'.
240 pieces at 14 EUR each sounds 'attractive' compared to
4 pieces of MJE13009...
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Old 26th September 2005, 11:35 PM   #30
Eva is offline Eva  Spain
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As opposed to low voltage ones, high voltage MOSFETs are quite expensive those days and prices don't seem to fall... rather they rise

Also, paralelling is almost always required for high power applications, thus multiplying the cost. I don't understand why engineers insist in using them everywhere, even in those designs that doesn't require extremely fast switching (like mine).
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