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31st October 2012, 02:32 AM  #1561 
diyAudio Member

Capacitance of electrolytics changes radically with frequency, and temperature.
Cornell Dubilier has a nifty visualizer and impedance modeler (provides frequencydependent/tempdependent SPICE models for their caps), at: Cornell Dubilier PlugIn Thermal / Life Calculator Attached is a screenshot for a particular 1000 uF electrolytic. If you design anything for low temperatures, you had better take a look. This one is fairly typical, in that sense. Last edited by gootee; 31st October 2012 at 02:36 AM. 
31st October 2012, 02:49 AM  #1562  
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You wouldn't want them sharing any conductor with your signal input ground, or any other lowleveltype signals or their grounds. 

31st October 2012, 03:12 AM  #1563  
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31st October 2012, 04:03 AM  #1564  
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From this perspective the decoupling cap should not be seen as a battery, rather as a snubber short circuiting high frequency noise ... Frank 

8th November 2012, 05:50 AM  #1565  
diyAudio Member

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Yeah. But I meant that the actual decoupling caps, which go from power rail to ground, very close to an active point of load, should never share a ground conductor with any signal ground or other lowlevel or sensitive ground. If they did, then their typically large and dynamic currents would probably induce relatively large voltages across the inductance and resistance of any shared ground conductors, making your nice quiet ground into a bouncing ground. And the bouncing ground voltage would then arithmetically sum with anything using it as a reference, which would be "a BAD thing". Decoupling caps go to power and load ground, which should only connect to signal and other quiet reference grounds back at the star ground point, so they don't share any length of conductor. For what you were talking about, I guess you would use separate caps, which wouldn't be called "decoupling" caps. The "decoupling" term means "decoupling a local power rail from the rest of the power rail network", in order to try to confine the large transient currents to a local loop, so as to not cause large voltage disturbances on other parts of the main power and ground rails due to trying to pull and push fastrising currents through their parasitic inductances and resistances. 

8th November 2012, 06:44 AM  #1566 
diyAudio Member

... been quiet lately, Tom ...??
Yes, the whole business of return and loop currents and grounds can become messy if one doesn't think it through, or use a tool like Spice. Rather than try and explain it a different way, I'll attempt to find an authoritive reference, or tutorial, somewhere on the net that has all the right diagrams on it, etc; give me a chance to find something good ... Frank 
8th November 2012, 08:41 AM  #1567 
diyAudio Member

Progress Update:
Just FYI, I am still working on a power supply spreadsheet (transformer, rectifier bridge, capacitor, and active load). I know that I said that it was almost done, way back when. But I found some problems and started a "deep dive" into it. Wow this sh!t is complicated and difficult. I think that I have well over half of it figured out, now. But implementing it has been a bit of a nightmare. However, if I can do it successfully, then it will probably be quite accurate. And it won't be only for finding Cmin. It would have been far easier to just have people use the LTSpice model that I have been using, especially now that we have the scalable transformer model, thanks to Terry Given, and AndrewT. But a spreadsheet will be far easier to use, for most people. However, had I realized how many times I would think that I was done and then realize that there was still a major discrepancy somewhere, that I didn't understand, I might have given up, long ago. (I'm trying to not think about the possibility that I am at a similar point, yet again.) One big revelation came, somewhat recently, when I saw that everything seemed to be looking very good EXCEPT where I was trying to account for the effects of transformer losses on the maximum rail voltage, due to the secondary resistance and the secondary and primary leakage inductances, during charging pulses. At that point I was mainly working with peak values of everything, including the peak charging pulse current (because the peak values of things were relatively easy to calculate and were mostly all that was needed). I apparently hadn't thought about it enough and was thinking that I could multiply the peak rectifier current by the sum of the transformer impedances to determine the voltage drop and then subtract that from the peak input voltage divided by the turns ratio to get the secondary's peak output voltage. Wow that was so wrong! Consider the charging pulse. A large swing of fastrising current and then a large swing of fastfalling current. It goes through a resistance and... an inductance, in the transformer model. This is not like AC analysis. Think "time domain", with a oneshot pulse through an RL series network. The voltage across the resistor follows the shape of the current pulse exactly, with a peak voltage determined by its resistance. So far so good. But the voltage across the inductor is V = L di/dt. And di/dt, the slope of the current waveform vs time, is first a large positive number and then a large negative number. So the inductance basically gets two oppositepolarity voltage pulses across it, during each charging pulse. And it doesn't end at zero, at the end of the pulse. And the total series voltage across the inductance plus the resistance is needed, to calculate what happens to the secondary output voltage. Actually, it's a little harder than that. The secondary "INPUT" voltage, at the output of the ideal transformer in the model (since the parasitics are separated out), during a charging pulse, is the theoretical secondary input voltage ("ideal" sinusoidal primary voltage divided by turns ratio) MINUS the voltage across the secondary leakage inductance. And, of course, VSEC_IN  VSEC_OUT = V_Ls + V_Rs, also. I noticed, while studying timedomain plots of the various voltages and currents from simulations, that at the beginning and end of a charging pulse, at least, VSEC_OUT = V_primary/turns_ratio 2 * V_Ls  V_Rs. But V_Rs is zero at the beginning and end of the pulse, since the current goes to zero there (but the slope of the current vs time doesn't go to zero there). So the delta_VSEC due to a pulse is Vprimary/turns_ratio (at pulse end)  2 * V_Ls (at pulse end)  Vprimary/turns_ratio (at pulse start). That's nice, except I didn't know how to calculate where the pulse start and end times were, in order to know where to pick the Vin values off of the input sine wave, so I'd be able to calculate the secondary's output voltage after a charging pulse (Hint: It can be wellabove the "ideal" theoretical maximum peak secondary voltage!). I had originally thought that VSEC_MAX was the theoretical peak secondary input voltage minus the voltage across the secondary leakage inductance at rectifier turnoff (pulse end), which would only require multiplying the final downslope rate of the pulse by the leakage inductance. That turned out to not always be correct. But I did learn to use Excel's curvefitting apparatus, and found a polynomial for the terminal slope of the charging pulses, under various conditions. But that equation changed for different VA and output power combinations. (So I started looking into converting sets of equations that give parallel plots in a plane into a moregeneral 2D "field" equation. But it turned out that "parameterization" is probably the usual best way to go, for that.) But then I found a better way to account for different VA and output power ratings, for that equation (mentioned farther below). I was able to plot all of the (simulation) voltages and currents that happen during a charging pulse, and had studied them so much that I finally deduced the simple equations showing which things add and subtract to produce the plots, which for some reason were just not obvious to me, at the beginning. But there seemed to be no easy way to calculate the actual values needed, in a spreadsheet, without an closedform mathematical equation for the charging pulse itself, i.e. some algebraic expression that would give current in amps if I plugged in a time value. The charging pulses somewhat resemble half of a rectified sine, in shape. But unlike a rectified sinusoid the beginning has a gradual starting slope at first, and the whole thing is asymmetrical and "slanted" and stretched toward the right, while the trailing edge's slope usually just gets steeper until it hits zero. (It turns out that, in my simulations at least, those righttrianglelike pulse shapes that some scientific papers' authors have used as an approximate model for the pulses are only produced when the parasitics are removed from the transformer model.) The charging pulses appeared to have the same basic shape and features, almost no matter what their peak values were. So I decided to "take the plunge" and measured about twenty or thirty data points from one, off of the LTSpice plot window screen, with my mouse cursor, and put the "measured" times and currents into Excel columns. Then I used the Excel curvefitting feature on that data and got a very good match by using a fourthorder polynomial. I measured two other ones, with different peak current values, too. I can get the maximum peak rectifier current, in my speadsheet, already, quite accurately, through other means. So I thought that it would probably be helpful (to say the least) if I could somehow "scale" the pulseshape equation from the one particular (16.36 Amp peak) rectifier pulse that I had an equation for, in order to be able to get an equation for a pulse that had any other peak value. So first I plotted, in Excel, the widths of the bottoms of several charging pulses, in milliseconds, versus their peak amps. Then I used the Excel curve fitter again and got a secondorder polynomial with zero error that will give the width of the bottom of a charging pulse (in milliseconds), given only its peak value. I then shifted the pulse data so that time=0 was at the peak and refitted the original equation, for that situation. I also normalized both the pulse widths and the peak values and redid that polynomial. That way, I could take a new peak value, calculate the width with the polynomial equation for that, and then use the original equation, but with the input time values simply scaled for the new width, and then scale the resulting current (amps) values by the new peak current divided by the one used for the original polynomial. It works! And it's trivial to differentiate and integrate polynomials. So now I also have equations, as functions of time relative to the peak's time, for the slope at any point in time, ih Amps per millisecond, and for the area under the curve between any two points in time, for almost any size of rectifier charging pulse (for the particular transformer model parameters supplied by AndrewT, at least). And note that the slope, for which I now have an equation, is the same "di/dt" that is needed to calculate the voltage across the secondary leakage inductance at any time. And the capacitor current's equations are now also known, since it's just the rectifier current minus the load current and my load current is a simple square wave. The rectifier pulse width also determines the exact time interval (or, equivalently, the phase angle range) for where the input sine wave gets "hacked into", during the charging pulses. So now, that simple equation that I gave earlier, for the "delta Vsecondary due to a charging pulse", will become very useful, as soon as I get one more step completed, which is calculating the exact time when the decaying capacitor voltage runs into the next rectified sine peak (or maybe the one after that, in some cases). It shouldn't be too difficult. It just happens to be the step I'm on right now, thanks to the fact that I now finally have a way to use that information to continue on and calculate the actual solution. It turns out that finding that intersection point, between the decaying exponential capacitor voltage and the rectified sinusoidal input voltage, can ONLY be done numerically (or graphically), since there is no closedform mathematical solution for a transcendental equation such as that. So it's a good application for a computer. I will (almost certainly) also have to go back one more time, at least, and redo all of the chargingpulserelated equations. Earlier, when I was playing around with curvefitting an equation for the maximum downslope at the end of a charging pulse, I eventually figured out that I could make it work for more than one output power and VA rating if I first multiplied the independent variable (the peak current value of a pulse for which the ending slope was desired) by the output power divided by the VA rating, and then curvefitted to find the equation. Then I could just first multiply any desired new peak charging pulse current value by Watts/VA, and then plug the result into the polynomial that had been found by Excel's curve fitter, and get the correct final slope for a charging pulse of any height, for any output power and transformer VA rating that was being used. So I assume that I will need to do something similar for the pulse_amps(t) polynomial and the related equations for its derivative (slope) and integral (area under curve). "So many fun things to do. But so little time." At any rate (at some rate?), I think that I will eventually get it done. Cheers, Tom Last edited by gootee; 8th November 2012 at 08:57 AM. 
8th November 2012, 09:49 AM  #1568 
diyAudio Member

Okay, Tom, I think I have the material that puts your point of view precisely: Ott's article "Ground  A Path for Current Flow". Here there are 4, yes, 4 separate grounds: power supply, amplifier, signal input, and load grounds for a highly conventional, single voltage, non feedback amp. All with significant impedance between them. But, the trouble with opamps is that there is no amplifier ground, no pin to which ground attaches, 2 supplies, and a very highly sensitive feedback network. In all, a different situation ...
So, I shall do some exploring to find, or work out, a "best" resolution ... Frank Last edited by fas42; 8th November 2012 at 09:52 AM. 
8th November 2012, 10:56 AM  #1569 
diyAudio Member
Join Date: Jan 2004
Location: Italy, Genova

Ciao my friends!
The JVC cdp is still on the surgery table... so no feedback, yet. Forgive me but I was simply thinking of placing a bunch of 330u/100V near the output transistors of a power amp. So, one leg to the "+" power rail, fine. The other leg to load common (i.e. speaker return) as per Ott's article (I found on http://www.electro.fisica.unlp.edu.a.../GROUNDS_2.pdf) should do the trick, correct? 
8th November 2012, 11:02 AM  #1570  
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Join Date: Aug 2008

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