Help on 2400W Phase Shift PSU

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Hi all,
I am designing a 2400W phase shift power supply and I have some problems that I don't understand.

The SMPS is a 24V 100A (150A peak current), it is connected to a buck-type
3phase PFC that I designed and that works perfectly (98.5% efficiency at 2500W out). The PFC stage delivers a 385V DC supply to the phase shift stage; the low voltage supply (VBSEC) is a 15Vdc supply derived from a small flyback SMPS connected directly to the rectified mains.

It seems I have a problem with the pulse transformer TR2. If the SMPS works in open loop (full duty cycle) everything is fine; as soon as it starts regulating
the output voltage the gate waveforms of Q12 and Q14 becomes DC biased (they are not symmetrical towards SGND). When this happens it seems that the synchronous rectifier goes crazy and close at the wrong time generating huge current spikes into the transformer. This behaviour is particulary critical between no load and around 5A load. The SMPS does not blow up but ugly sounds are generated by the transformer and the primary current hits the current limit point set at 18Apeak.

I think that the problem is located in the gate driver; I have already tried to change the gate resistors and removing the secondary snubbers (C26-R34) with no success.

I have found on Ti website an app note regading the asymmetry of outc-outd during duty cycle variations. I have applied the fix described in the app-note with no success.

Does anyone of you have experience with phase shift SMPS and in particular with UCC3895 controller?

If anyone has useful ideas to share with me it will be highly appreciated.

Thank you

-marco

Other useful data:

Transformer TR1: EE55 core no gap, primary 21turns, secondary 4turns
Inductor L4:EE25 core with ~1mm gap, 8turns
Pulse transformers TR2,TR3: 10turns, trifilar wound on 16mm ferrite core
Current doubler: L5, L6: 13turns on kool-mu 50mm toroidal core
 

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Sensing current bidirectionally that way with a single transformer does not work, it results in asymmetrical operation because the current transformer can't sense DC errors (it will tend to saturate in the direction of the DC error). So any DC error will not only impair AC sensing but DC sensing too (by sensing less current in the direction more is flowing.

Use either 2 transformers (one for each direction), one transformer unidirectionally with duty cycle limiting for reset, or shunts with a pole on sensed voltage drop for parasitic inductance compensation.
 
What Eva said. I think one of the earlier 3875s actually had that cs xfmr in series with the main transformer. Won't work.

You can use 1 transformer and measure the current in FET A and B drain. At full duty cycle there's almost no reset time so voltage spikes can be quite large.

You can also use 2 xfmrs, one in the FET A source and the other in FET B drain and sum them on the secondary side.

I've used both methods in converters with good sucess

You can also get rid of the caps in series with the power xfmr.
 
Hi all,
in fact without the series blocking capacitor the trasformer saturates due to DC. Theoretically using current mode control this capacitor is not required but in my case it is absolutely needed.

@switchmodepower

In fact this current transformer connection comes from an app note of the old
UC3875, the new designs with 3895 connects the current transformer between the two mosfet drains and the power supply input.

In my design the operating duty cycle is around 65% and to connect the current tranasformer in this way I must be sure to fully demagnetize it in the
interval when both upper mosfets are off, otherwise it will saturate.

I have tried to disable the synchronous rectifer (I removed the drive signal and I use only the mosfets body diodes) and the SMPS at light load still make nasty sounds from the transformer, the drive signal of C and D mosfet is sill not symmetrical wrt. to GND. In those conditions the DC voltage across the transformer DC blocking capacitor reach 40-50Vdc; when the load is increased the DC voltage becomes much lower (<5V).

I think that my brigde is unbalanced but the question is:
is the bridge unbalanced because of drive issues or it is unbalanced because of a wrong current sense signal due to current trasformer position?

Thank you
 
What about running the UCC3895 in voltage mode instead of peak current mode control?

I know that there will be for sure an imbalance in the brigde that creates a DC offset, but that offset will be cancelled by the coupling capacitor.

The output current will not for sure exacly share at 50% over the 2 current
doubler inductances due to imbalance but is it so important?

I have seen some industrial applications running the UCC3895 in voltage mode with a current doubler rectifier and they works ok.

If I run in voltage mode I can even leave the current transformer as it is now since it will be used only for peak current limiting and for adaptive delay set (not so critical). Changing the position of the current transformer is very difficult for me on the current layout and I prefer to avoid it if possible.

Thank you

-marco
 
Awww. Don't give up. Current mode works great with this topology...something is wrong. FIND IT!!

Take the series caps out. It will never work in CM with them.

What does the ramp signal look like at the chip? This signal cannot be currupted at all. Make sure that you have single point connections from the cs xfmr to the chip grnd so that you;re not picking up noise.

You said that you see drive signals that are not symetrical at no load. This is an indication of the 'save me' CS trip. If you isolated the CS pin and short it for an experiment does this go away or blow up?

You can also add more slope to help at the lower loads. Not too much or you'll be operating closer to voltage mode.

Good idea about getting rid of the sync FETS for now. Strip out anything that can cause problems and just worry about what wrong or you can be chasing your tail.
 
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Also. Have you just tested the power stage? I know you said you ran in open loop but what about open voltage loop but closed current loop?

Get rid of the voltage op amp and feed in a DC signal into the EAP of the chip. A potentiameter tied to 5V ref works good. Consider this your current command signal. Set your load to full load and adjust the pot from zero voltages upward.

You should now be able to operate the converter in a current source fashion with the current set by the pot. The output voltage will vary.

If you can get this mode to work you're on a good track and most likely it's a loop thing.. If this does not work then there's really something wron.
 
Hi switchmodepower,

I have tried to check the current loop using the current error amplifier.
If I put a direct short on the output the behaviour is quite unstable with the
CS 'save me' tripping sometimes. If instead of a short I put a very small resistor (around 0.1ohm) everything looks fine and I can regulate the output current.

Sorry I did not tell you that there is another op amp connected to VFB that closes the loop using the information from the output current hall sensor to regulate the output current. This is not for sure the root of the problem because even if I disconnect it the behavior does not change.

The signal on the RAMP pin is quite bad, especially at low load when the ramp
coming from the CS transformer is not well defined. I tried to increase the amount of slope compensation but it does not help.

I tried also tu put the chip in voltage mode and everything seems ok: no more instabilities at low load, no nasty sounds and no brigde imbalance.
I have mesured the DC voltage across the transfomer series cap and it remains below 0.2V from no load to around 50A load.

Now I am supplying the converter with a 200V 1000W laboratory power supply (I am not using the PFC) and I have regulated the output at around
12V instead of 24V. I have done this because if something goes wrong I have the 200V power supply current limit that saves me and I avoid to blow up everything...

I have not tried to short the CS and see what happens; if t blow up in a bad way with such power there is the risk to burn the PCB traces and I have only one prototype available.
Btw: what you call the 'Save me' protection of the CS means the second trip point at 2.5V or other?

I think now that the problem is in the current transformer; good hint to route the transformer GND directy to the chip GND, now it is connected on a common GND plane on the PCB and there is the possibility of noise pick up.


Removing the series cap is not possible, if I try this there are some loads (not only a low load but also at heavy loads) where the transformer saturates and the CS trips in.

Going in voltage mode seems to solve all my problems, so why don't go for it? What are the possible problems?

When in voltage mode I have also tried to reconnect the synchronous rectifier and everything goes well. The only problem is that when the mosfets body diodes conduct there is a huge ringing voltage spike on the output mosfets drains; an RC snubber is for sure needed here.

Next week I will do some new test changing the position of the current transformer (between the positive rail and the upper mosfet drains) using current mode control with no series cap. But before doing this I need to damp the ringing on the syncronous rectifier, now with 200V input the spikes reaches around 70V and the SR mosfets are rated 100V. When I power up with the PFC (385V) those spikes can reach the mosfets BVDss.

Thank you for your support
 
Yes, I have still the xfrm in series with the primary.
I will change its position before trying to run in current mode.

I will put it in series with the two upper mosfets drains.

But before doing this I need to add snubbers on the output rectifiers to damp
the ringing

thank you
 
Hi all,
I have made some experiments with voltage mode and everything seems good. The current xfrm is still in series with the primary and I am using the DC block capacitor.

I have pulled 80A / 24V from it with a global efficiency (PFC+converter) of 93.5%. At 80A there are some PCB track that starts to heat a lot, I need to reinforce them before going to 100A.

One question: my SMPS will work also in constant current mode (with a separate error amplifier). If I connect to a resistive load of around 0.1ohm I can easily regulate the current from 10A to 80A with no problem. If I connect a real short circuit (short 25mm^2 wire directly on output) there are some instabilities. I am able to regulate the current in the short but sometimes the primary current limit kicks in. This happens also if I set the output current to around 10A.

Is this happening because I am tring to operate a voltage mode converter as a constant current source or can it be just a loop compensation problem.

After changing to voltage mode I have changed my volage loop to type 3 with around 6kHz BW and 70° phase margin. What about the current loop?
No idea on how to proper compensate the converter when it runs as a constant current source, do I need type 3 or type 2 can do the job?
Any suggestion will be appreciated.

Thank you

-marco
 
Hi all,
I have made some experiments with voltage mode and everything seems good. The current xfrm is still in series with the primary and I am using the DC block capacitor.

I have pulled 80A / 24V from it with a global efficiency (PFC+converter) of 93.5%. At 80A there are some PCB track that starts to heat a lot, I need to reinforce them before going to 100A.

One question: my SMPS will work also in constant current mode (with a separate error amplifier). If I connect to a resistive load of around 0.1ohm I can easily regulate the current from 10A to 80A with no problem. If I connect a real short circuit (short 25mm^2 wire directly on output) there are some instabilities. I am able to regulate the current in the short but sometimes the primary current limit kicks in. This happens also if I set the output current to around 10A.

Is this happening because I am tring to operate a voltage mode converter as a constant current source or can it be just a loop compensation problem.

After changing to voltage mode I have changed my volage loop to type 3 with around 6kHz BW and 70° phase margin. What about the current loop?
No idea on how to proper compensate the converter when it runs as a constant current source, do I need type 3 or type 2 can do the job?
Any suggestion will be appreciated.

Thank you

-marco

hello Marco,

any chance of posting the PFC schematic?
i am about to start work at a 5kw PFC and anything already made at kw levels would be of great info for me.

regards,
 
hello Marco,

have you already solved your problems? I'm actually also designing a similar power supply and possible we can share some experiences.

regards

marco, may this work for current sensing instead of using many sensor at each mosfet leads.

yes please continue the post. and the pfc sch too:)
I'm now designing buck based PFC for 500W limited load with 150V stable output.

May I asking why using phase shift and why full bridge instead of half?
thanks
 

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With average current mode and a plain short circuit load, find out why current is increasing over the limit, how and when. You will need a storage oscilloscope and careful triggering to capture the waveforms.

It may be transformer saturation or overshoot due to loop instability.

Note that a plain short circuit load will turn output capacitors into a short, thus turning output filter LC, a 2nd order system, into just L driving a short to ground, a 1st order system with lower gain. Unity gain frequency will change. But I think the problem is transformer saturation (due to current transformer saturation).
 
With average current mode and a plain short circuit load, find out why current is increasing over the limit, how and when. You will need a storage oscilloscope and careful triggering to capture the waveforms.

It may be transformer saturation or overshoot due to loop instability

The output voltage may just too low, or may be transformer.
Low output voltage will cause the inductor (transformer in this case) flux not discharged (look this graph).
In current mode also the transformer designed different from voltage mode. In picture above, there four inductor in current mode power transfer, I placed small ferrite in center with paper as spacer. It will have better flux density than without spacer. Please take a time for looking current mode (flyback) transformer design.

Hi Eva, how your class-D?
 

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@savu.

sorry I can not share the PFC schematic but I can tell you more or less how I did it.

I have used a borderline PFC controller to control a buck converter. The mosfet is driven through an optoisolated drive. The buck converter input is
connected to the output of a 3-phase bridge rectifier with some L-C filters to reduce input ripple current and thus EMI content.
At the output of the buck converter I have the capacitors bank and I regulate the output at around 385V.
The PFC works always in DCM or in BCM so there are no recovery time issues for the diode. Working in BCM allows also to reach quasi ZVS operation.

The PFC is designed for an input voltage between 350 and 500Vac 3phase.
The PF is around 0.95 and the input current is not a sinewave but has far less harmonic content than a standard diode-capacitor input stage so that I can fullfill class-A requirements on line harmonics.

@onto aban

I can not use a shunt resistor for current sensing; I need isolation. I have my
controller on the secondary side and I must sense the current on the primary side, a current transfomer is the easiest way. The best way to do the job with resistors is to connect the lower side mosfet sources toghether and then connect them to GND through a sense resistor. In this way you have your current sense volatge always referred to GND.

This type of PFC works only for 3phase systems. If you need 150V 500W output from your PFC and your input is a single phase AC this PFC will never
work. This because the buck goes at 100% duty cycle when the input AC falls below 150V and it will not correct the PF.
The good think of 3phase rectifier is that the voltage never goes to 0V, the
rectifier ouptut is an haversine of 300Hz with the maximum at Vin*sqrt(2)
(565V in a 400VAC system) and the minimum at Vin*sqrt(2)*sin(60°)
(490V in a 400VAC system), the point where two phases crosses. In my case
regulating the output at 385V ensures that the buck is always working because Vout < Vin at every moment of the line cycle.

If you need 150V stable a PFC of every kind is not the way to go. The PFC
output has many volts of 2*fline ripple and this is normal; more ripple you allow the better the PF will be.

If your system is single phase and you need a 150V 500W stable output for me there are only two way to go:

1) standard boost PFC (possibly CCM) regulated at around 400V +
buck regulator to step down the 400V to 150V

2) Flyback PFC, it removes the limitation of having vout>vin of the classical boost PFC. You have a transformer so you can play with the turn ratio to have the output voltage you need. It give also galvanic isolation if you need it. In any case your output voltage will not be a pure DC at 150V but it will have a 100Hz ripple of some volts. If you can accept this I think that this is best way to go.

I have used the phase shift topology because it allows a simple drive of the synchrounous rectifier and using the current doubler I can reduce the current
in the main tranformer from 100A to 50A (more or less) an the primary switches can switch at zero voltage elimitating switching losses. The expense is two more mosfet on the primary side and a more complicated controller and gate driver.



@ Eva

You got exacly the problem. With a short on the output the system becomes of 1st order and there are no way to compensate it with an adeguate phase margin using a standard type 3.
If you tune the compensator for the plain short (1st order) it will be unstable
when the output voltage rise and the system is operating as a current source (2nd order) and vice versa.
I have developed an alternative compensation network adding other two zeros on the compensators to have less phase boost but over a wider bandwidth. With this compensator I have always around 45 degrees of phase margin sweeping the load resistance from 1mohm to 200mohm.
My problem is now fixed and the system is stable on every load even when using it as a current source.
Now I am still using voltage mode, with series cap and with the current transformer in series with the primary of the main transformer. Nothing seems to saturate and everything runs fine and cool except the inductance in series with the primary (wrong calculation of core losses--> I need a bigger core with more turns).
At this stage I don't think to move the current transformer to use current mode control, everything is fine in voltage mode so why change it?

@onto aban
This is a forward converter, I need to reset the transormer only for its magnetizing current, not the full current as in the flyback.
 
@savu.

The PFC is designed for an input voltage between 350 and 500Vac 3phase.

Industrial?



@savu.

I can not use a shunt resistor for current sensing; I need isolation.

Transformer still there, but high impedance (voltage mode) calculated from R voltage.



@savu.
If you need 150V 500W output from your PFC and your input is a single phase AC this PFC will never
work. This because the buck goes at 100% duty cycle when the input AC falls below 150V and it will not correct the PF.

Yes, that's right. Its already three converter. Two of them working at low power. The output is stable 150V but I am still have big problem with this.
Using flyback need big electrolyt caps, I just avoid this one.


@savu.

I have used the phase shift topology because it allows a simple drive of the synchrounous rectifier and using the current doubler I can reduce the current
in the main tranformer from 100A to 50A (more or less) an the primary switches can switch at zero voltage elimitating switching losses. The expense is two more mosfet on the primary side and a more complicated controller and gate driver.

Interesting, is the transformer size reduced?

@savu.

My problem is now fixed and the system is stable on every load even when using it as a current source.
You mean short circuit at the output? My amp did it, still operating at short output, so there no need recovery when the short opened. I simply place impedance limiter from the current sense to limit the input and let the output devices dissipate low power. Since it is class-H the amp working at lowest rail.
You are using switching that none will dissipate any, I have no idea.
 
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