A headphone Transconductance Amp for a change!
The attached schematic shows one channel of an OP Amp driving a buffer pair of complementary output transistors (TIP 120, and TIP 125 Darlingtons). This output stage is operating in the common emitter rather than the customary common collector configuration. Thus, it is an inverting amplifier with voltage gain if desired. This transconductance or current source amp requires 2 separate power supplies for proper operation. The schematic shows one +/- 6V supply to the OP Amp and another higher voltage +/-15 V for the output stage. The output of this amplifier is the common port or center point of the +/- 15V supply. Thus the headphones are connected between this port and ground. By ground I mean the center point of the +/-6V supply. The prototype was assembled on an experimenter's proto board. The attached images are not pretty; but they confirmed its feasibility. One image is a top view and the second is taken from the side of the volume control. The components are discernible. The above power supplies I used happen to be (ginormous) surplus from spent computers.
This amp is capable of a high outupt current into headphones. Note the resistor of 220 K which attenuates the input signal so as protect one's hearing. The quiescent conditions of the amp are:
- 10 mV dc offset at output port
- 5 mVpp noise/hum at output port
- Output transistors idle at ~23 mA each
- Op Amp is RC 4560 which sounded slightly better than OPA2134
- Must have output Zobel, a 10 pF ceramic cap (between inverting input and output of Op Amp) and the 100 K feedback resistors to prevent oscillation
- Headphone was Grado SR 80 connected in parallel (mono). It must be connected before turn on so as to complete the circuit of the output stage.
I believe DIYers (and I) will run with this "square one" amplifier and tweak it so as assemble in the longer range the best there will be.
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Post moved to new thread, please do not post unrelated projects to the O2 thread.
Have you checked to see whether the pdf is too large? Break it into a number of smaller pdfs - if converting from jpg reduce the color depth or even make it black and white, increase the compression levels before converting to pdf.
Edit: I see you succeeded.
It is a fact of life. Some of us DIYers are averse to using IC Op Amps in the path of a music signal. Our fellow DYIers have a highly critical and discriminating sense of hearing. One fall back to avoiding such signal IC Op Amps is to use Op Amps which are assembled from discrete components. A power amp is an example of a discrete albeit powerful Op Amp which has the capability to deliver both a high voltage and a high current into a loudspeaker (power). Music signals pass right through and out of it! A second fall back is to use IC Op Amps in a role which "supports" an amplifier which is neither an IC nor a discrete Op Amp. An example is a servo DC amp which is used around the signal amplifier so as to primarily remove a DC offset from its output port; for example as it drives headphones. This supportive IC Op Amp may also need to be a high performer e.g. OPA2134 or RC4560. The signal amplifier needs the highest quality support it can get. The last fall back is not to use either an IC or a discrete Op Amp at all. This approach presents its own design challenges and trade offs.
In the preceeding thread(s) , I have shown the schematic of a working Headphone TransConductance Amplifier (HTCA) which used a high quality IC Op Amp in the path of the musical path. Some fellow DYIers may have tuned out in a spontaneous gesture of disapproval! In an upcoming thread, I will show two schematics of a Class A (HTCA) which is based on a discrete Diamond Buffer for processing music signal and also embodies:
In the preceeding thread(s) , I have shown the schematic of a working Headphone TransConductance Amplifier (HTCA) which used a high quality IC Op Amp in the path of the musical path. Some fellow DYIers may have tuned out in a spontaneous gesture of disapproval! In an upcoming thread, I will show two schematics of a Class A (HTCA) which is based on a discrete Diamond Buffer for processing music signal and also embodies:
- A high quality IC Op Amp as a DC servo Amp to null DC offset voltage at its output
- No IC or discrete Op Amp in any role therein
A headphone Transconductance amp based on a Diamond Buffer
The attached pdf: Headphone 003 is a schematic of a refined Headphone Transconductance amp. It does not utilize IC OP Amps in the path of the music signal and around it. It is based on the Diamond Buffer; popularized in its hay day by National Semiconductor. The input signal Vi is presented to the bases of complemetary symmetry transistors 2N3440 (NPN) and 2N5415 (PNP). These transistors are working in the common collector configuation; meaning the buffered signal is extracted from their emitter ports. Lets focus on the positive half of the circuit. This begins with 2N5415 (PNP) and is traced all the way up to the +15 V supply rail. The discussion applies fully to the bottom half which begins with 2N3440 (NPN) and is traced to the -15 V supply rail due to complementary symmetry. Notice that I labelled only the values of the circuit elements for the upper half. And I labelled the equivalent circuit elements of the bottom half with the quiescent voltage drops and currents passing through them. These quiescent conditions for the upper half are identical to the lower half due to complementary symmetry. This gives the broadest picture of the schematic
In the ideal world of semiconductors, the front end transistors are highly matched. Thus, the base current entering the NPN transistor is equal in value to that leaving the base of the PNP complement. They cancel to a net zero current passing through the input resistor (47K) from the ground node. This resistor can be as high as 1Meg, and thus there is no input offset voltage. In the real world; the base currents of the input transistors are mismatched, and did cause a non-zero input offset voltage in the actual circuit (+0.712 V). The circuit shown in the bottom left of the schematic is used to null as best as possible this input offset, and it simulataneously nulled the voltage offset at the output node of the amplifier. Piece of cake to implement! Still I used a DC coupling capacitor in the input circuit to block the residual offset voltage (- 26 mV) so as to protect the output source of the input signal.
The upper N-channel JFET (2N3819) sources or (sinks) a constant current (~3.1 mA). This current flows through the red LED to cause a voltage drop of (1.875V), and then through the Si diode to cause an additional voltage drop of (0.68 V), and then through the body of 2N5415 to cause an additional voltage drop Vbe of (0.68 V). The constant current finally exits the 2N5415 transistor out of the collector (mostly) and base (trivial) ports. The sum of the above diode voltage drops is equal to (3.24 V). This is the bias voltage for the NPN Darlington Transistor TIP 120 and its emitter resistor (30 Ohm). The calculated emitter current is equal to 65 mA. This transistor is working in Class A and in the common emitter configuration. Because a lot of current is flowing through it and its emitter port goes to ground via the reistor (30 Ohm). This NPN TIP 120 is powered by a separate +6 V supply; modelled as a battery. In its operation, its emitter current Ie is equal to its collector current Ic; by neglecting the trivial current entering the base port and adding to Ie. So, current Ic begins to flow from the positive terminal of the upper 6 V battery, enters the collector port, passes through the physical body of the transistor, exits the emitter port, passes through the resistor (30 Ohm), does not stop at ground, but continues its journey through the load resistor (15 Ohm = headphones) and finally reaches the negative terminal of the upper 6 V battery where it belongs (clearly exhausted after all this work).
The impedance at the output port [Vo] is high because of the opposed collector connection of the TIP transistors. The batteries are dead shorts for a music signal. This output impedance can be several hundred Ohms or more. The "product" of this output stage is current and not voltage. Thus it is called a current source amp or a transconductance amp.
I chose a +/-6 V power supply for the output stage because it is fully adequate. The oscilloscope trace registered at best 200 mV peak to peak music signals across the 15 Ohm headphones I used. Protect your ears.
The attached pdf: Headphone 003 is a schematic of a refined Headphone Transconductance amp. It does not utilize IC OP Amps in the path of the music signal and around it. It is based on the Diamond Buffer; popularized in its hay day by National Semiconductor. The input signal Vi is presented to the bases of complemetary symmetry transistors 2N3440 (NPN) and 2N5415 (PNP). These transistors are working in the common collector configuation; meaning the buffered signal is extracted from their emitter ports. Lets focus on the positive half of the circuit. This begins with 2N5415 (PNP) and is traced all the way up to the +15 V supply rail. The discussion applies fully to the bottom half which begins with 2N3440 (NPN) and is traced to the -15 V supply rail due to complementary symmetry. Notice that I labelled only the values of the circuit elements for the upper half. And I labelled the equivalent circuit elements of the bottom half with the quiescent voltage drops and currents passing through them. These quiescent conditions for the upper half are identical to the lower half due to complementary symmetry. This gives the broadest picture of the schematic
In the ideal world of semiconductors, the front end transistors are highly matched. Thus, the base current entering the NPN transistor is equal in value to that leaving the base of the PNP complement. They cancel to a net zero current passing through the input resistor (47K) from the ground node. This resistor can be as high as 1Meg, and thus there is no input offset voltage. In the real world; the base currents of the input transistors are mismatched, and did cause a non-zero input offset voltage in the actual circuit (+0.712 V). The circuit shown in the bottom left of the schematic is used to null as best as possible this input offset, and it simulataneously nulled the voltage offset at the output node of the amplifier. Piece of cake to implement! Still I used a DC coupling capacitor in the input circuit to block the residual offset voltage (- 26 mV) so as to protect the output source of the input signal.
The upper N-channel JFET (2N3819) sources or (sinks) a constant current (~3.1 mA). This current flows through the red LED to cause a voltage drop of (1.875V), and then through the Si diode to cause an additional voltage drop of (0.68 V), and then through the body of 2N5415 to cause an additional voltage drop Vbe of (0.68 V). The constant current finally exits the 2N5415 transistor out of the collector (mostly) and base (trivial) ports. The sum of the above diode voltage drops is equal to (3.24 V). This is the bias voltage for the NPN Darlington Transistor TIP 120 and its emitter resistor (30 Ohm). The calculated emitter current is equal to 65 mA. This transistor is working in Class A and in the common emitter configuration. Because a lot of current is flowing through it and its emitter port goes to ground via the reistor (30 Ohm). This NPN TIP 120 is powered by a separate +6 V supply; modelled as a battery. In its operation, its emitter current Ie is equal to its collector current Ic; by neglecting the trivial current entering the base port and adding to Ie. So, current Ic begins to flow from the positive terminal of the upper 6 V battery, enters the collector port, passes through the physical body of the transistor, exits the emitter port, passes through the resistor (30 Ohm), does not stop at ground, but continues its journey through the load resistor (15 Ohm = headphones) and finally reaches the negative terminal of the upper 6 V battery where it belongs (clearly exhausted after all this work).
The impedance at the output port [Vo] is high because of the opposed collector connection of the TIP transistors. The batteries are dead shorts for a music signal. This output impedance can be several hundred Ohms or more. The "product" of this output stage is current and not voltage. Thus it is called a current source amp or a transconductance amp.
I chose a +/-6 V power supply for the output stage because it is fully adequate. The oscilloscope trace registered at best 200 mV peak to peak music signals across the 15 Ohm headphones I used. Protect your ears.
A headphone Transconductance Amp based on a Diamond Buffer
It is important to determine the DC voltage offsets which appear at both the input [Vi] and output [Vo] ports of the amplifier. These voltage offsets are a direct consequence of mismatches in the characteristics of the complementary devices and differences in the voltage drops across diodes and resistors used therein the circuit. An excessive level of DC voltage at the output relative to ground and across the headphone may damage it and/or move its diaphragm off its center or midpoint. This limits the excursion of the diaphragm one way or the other so as to possibly cause distortion. At the input port, one needs to know the polarity of the voltage relative to ground so as to properly connect a DC blocking electrolytic capacitor (if needed) between the input port and the music signal source.
The attached schematic PDF Headphone004 shows the circuit which was used to determine the DC voltage offsets at [Vi] and [Vo]. Note the following:
Table 2 shows the DC voltage offset results with the input switch in the ON position. The trend of [Vo] as a function of [Vi] is exactly like shown in Table 1. The Input Impedance of the amplifier is now [R = 15 K Ohm] which gives a tolerable [Vo ~ 50 mV] offset. The connection of [R] between the input and output ports gives its circuit a weak DC Servo reaction; because the amplifier is inherently phase inverting. Resistor [R] appears as a feedback resistor; clearly from [Vo] back to [Vi]. However, the DC servo reaction and this global loop feedback are trivial compared with those enabled by an Operational Amplifier for comparison. The reason is simple, the indicated transconductance amp does not have an open loop (R = infinite, input switch is off] or a closed loop [input switch is on] voltage gain to speak of; like the many hundreds of an Op Amp.
I prefer the circuit with the input switch ON and thus R = 15 K Ohm [Table 2]. It has the mild protection of the DC servo seesaw effect.
Listening to the amplifier. I have a SONY CDP-C515 compact disc player. It has a remote volume control for its output RCA port and simultaneously for its internal headphone output. The output RCA ports need to look into an input source impedance which is greater than 10 K Ohm; which is the case here. I used either the Right Or the Left RCA outputs. Merging the 2 output RCA ports with a Y connector to make a mono signal is doable; but it did not sound well. The headphones were Grado SR-80 connected in mono (R = 16 Ohm). They sounded detailed, and tonally balanced.
The headphone output on the SONY is suspected to be driven by an IC Op Amp working as buffer of a low output impedance. Ironically, I used it as the model for the low impedance source of the music signal to drive the headphone transconductance amplifier. I used a conversion adapter from a stereo 1/4 inch plug to a stereo female RCA. Like above, I used either R or L ports, and did not merge the ports via a Y connector to make a mono signal. Again, the headphones sounded detailed and great. Maybe slightly different in tonal balance relative to the above case.
It is important to determine the DC voltage offsets which appear at both the input [Vi] and output [Vo] ports of the amplifier. These voltage offsets are a direct consequence of mismatches in the characteristics of the complementary devices and differences in the voltage drops across diodes and resistors used therein the circuit. An excessive level of DC voltage at the output relative to ground and across the headphone may damage it and/or move its diaphragm off its center or midpoint. This limits the excursion of the diaphragm one way or the other so as to possibly cause distortion. At the input port, one needs to know the polarity of the voltage relative to ground so as to properly connect a DC blocking electrolytic capacitor (if needed) between the input port and the music signal source.
The attached schematic PDF Headphone004 shows the circuit which was used to determine the DC voltage offsets at [Vi] and [Vo]. Note the following:
- The triangle symbolizes the amplifier circuit so as to simplify the schematic and thus make it clear (less congested). This triangle implicitely includes all of the circuit elements which were used in PDF Headphone003 (previous thread) like the transistors, resistors, diodes and power supplies.
- Use a ZOBEL at [Vo] to minimize ultrasonic broadband noise.
- The switch at the input port [Vi] presents two options. The OFF position connects the variable input resistor [R] to ground. The value of this input resistor is the Input Impedance which the amplifier presents to the music signal source. The ON position of the switch connects the input resistor [R] to the outupt node [Vo]. The Input Impedance of the amplifier is still equal to the numerical value of [R].
Table 2 shows the DC voltage offset results with the input switch in the ON position. The trend of [Vo] as a function of [Vi] is exactly like shown in Table 1. The Input Impedance of the amplifier is now [R = 15 K Ohm] which gives a tolerable [Vo ~ 50 mV] offset. The connection of [R] between the input and output ports gives its circuit a weak DC Servo reaction; because the amplifier is inherently phase inverting. Resistor [R] appears as a feedback resistor; clearly from [Vo] back to [Vi]. However, the DC servo reaction and this global loop feedback are trivial compared with those enabled by an Operational Amplifier for comparison. The reason is simple, the indicated transconductance amp does not have an open loop (R = infinite, input switch is off] or a closed loop [input switch is on] voltage gain to speak of; like the many hundreds of an Op Amp.
I prefer the circuit with the input switch ON and thus R = 15 K Ohm [Table 2]. It has the mild protection of the DC servo seesaw effect.
Listening to the amplifier. I have a SONY CDP-C515 compact disc player. It has a remote volume control for its output RCA port and simultaneously for its internal headphone output. The output RCA ports need to look into an input source impedance which is greater than 10 K Ohm; which is the case here. I used either the Right Or the Left RCA outputs. Merging the 2 output RCA ports with a Y connector to make a mono signal is doable; but it did not sound well. The headphones were Grado SR-80 connected in mono (R = 16 Ohm). They sounded detailed, and tonally balanced.
The headphone output on the SONY is suspected to be driven by an IC Op Amp working as buffer of a low output impedance. Ironically, I used it as the model for the low impedance source of the music signal to drive the headphone transconductance amplifier. I used a conversion adapter from a stereo 1/4 inch plug to a stereo female RCA. Like above, I used either R or L ports, and did not merge the ports via a Y connector to make a mono signal. Again, the headphones sounded detailed and great. Maybe slightly different in tonal balance relative to the above case.
Simplified the headphone Transconductance Amp.
There is French proverb which translates to: "simplicity makes beauty". The simplified transconductance amp is shown in PDF Headphone005. Three objectives were sought and met. First, the input complementary symmetry transistors 2N3440 (NPN) and 2N5415 were removed. They were replaced by the 2 diodes labelled 1N914* so as to establish the original bias of the output transistors. Second, the input resistor was raised to 47K; why dump valuable input signal to ground by using a smaller one like 15 K?. This resistor can be either grounded or connected to the output port Vo. This latter connection is a mild DC servo and is preferred. It maintains the DC offset at the output port ~ zero volts. Thirdly, the resultant DC offset at Vo slowly cycled over several minutes (due to servo action) between + 15 mV and -15 mV. Note the resistor (267 K) which is connected between minus 15 V and the input port Vi. Without it, the DC offset at Vo before servo action was about minus 1.3 VDC; this is unacceptibly high. With it and without DC servo (i.e. 47 K grounded) the offset at Vo came down to ~50 mV. The value of this resistor was arrived at by trial and error. Start with 1 Meg Ohm and lower its value by sequentially using 680K, 470K, 330K etc until a target 50 mVDC is obtained at Vo. I show a connection to minus 15 VDC. In the hands of another DIYer the connection may need to be to plus 15 VDC instead; because of different element mismatches in our respective prototypes. This modified amp sounded more relaxed than earlier versions.
There is French proverb which translates to: "simplicity makes beauty". The simplified transconductance amp is shown in PDF Headphone005. Three objectives were sought and met. First, the input complementary symmetry transistors 2N3440 (NPN) and 2N5415 were removed. They were replaced by the 2 diodes labelled 1N914* so as to establish the original bias of the output transistors. Second, the input resistor was raised to 47K; why dump valuable input signal to ground by using a smaller one like 15 K?. This resistor can be either grounded or connected to the output port Vo. This latter connection is a mild DC servo and is preferred. It maintains the DC offset at the output port ~ zero volts. Thirdly, the resultant DC offset at Vo slowly cycled over several minutes (due to servo action) between + 15 mV and -15 mV. Note the resistor (267 K) which is connected between minus 15 V and the input port Vi. Without it, the DC offset at Vo before servo action was about minus 1.3 VDC; this is unacceptibly high. With it and without DC servo (i.e. 47 K grounded) the offset at Vo came down to ~50 mV. The value of this resistor was arrived at by trial and error. Start with 1 Meg Ohm and lower its value by sequentially using 680K, 470K, 330K etc until a target 50 mVDC is obtained at Vo. I show a connection to minus 15 VDC. In the hands of another DIYer the connection may need to be to plus 15 VDC instead; because of different element mismatches in our respective prototypes. This modified amp sounded more relaxed than earlier versions.
Simpler headphone Transconductance Amp
The schematics in the the past threads showed the use of TWO Power supplies to drive the amp. No doubt a turn off to the DIYer. Do we need two power supplies?. Absolutely not. The past shematics also showed using a regulated power supply. Another turn-off! I had it at hand and it was convenient to use. Fortunately we do not need a regulated power supply to drive the amp. We only need one unregulated [dual] power supply per channel in the range of +/-10 VDC to +/-20 VDC. Figure 1 in the attached PDF Headphone 006 shows the [even] simpler schematic of the transconductance amp. It is mostly like that shown in the previous thread less the now-obsolete +/- 6 VDC supply. The schematic also shows the use of one dual +/- 15VDC power supply. The schematic of this common power supply is shown in Figure 2. It is comprised of a power transformer (12Vac secondary plus a center tap), a bridge rectifier and 2 filter capacitors (4700 microfarad each). Note its the battery model which I used in Figure 1. The CENTER TAP of the transformer's secondary joins the common junction of the filter capacitors, and most importantly it is the OUTPUT PORT Vo and not the customary ground.
In operation, the variation in the output voltage Vo is a trivial disturbance to the working of the JFET constant current sources (CCSs). The +/- 200 mV peak to peak level of Vo across the headphone [for my comfortable listening] is less than 1% voltage variation across the CCSs; i.e. (0.2V signal divided by 30 V power supply times 100 = 0.7%). Also, the value of the current passing through the JFETs is quite stable over a wide range of power supplies. Thus the JFETs resist the voltage variations in Vo and present themselves to the power supply as impedances of high value.
Food for thought.
Suppose I use two resistors (5.6 K each instead of the JFET contant current sources. One resistor is connected to the +15 V and the other to -15 V. The current passing through them is 30 V divided by 11.2 K = ~2.7 mA which also flows through the diodes to bias the output transistors. In operation the situation is quite different from the one I described above.
The schematics in the the past threads showed the use of TWO Power supplies to drive the amp. No doubt a turn off to the DIYer. Do we need two power supplies?. Absolutely not. The past shematics also showed using a regulated power supply. Another turn-off! I had it at hand and it was convenient to use. Fortunately we do not need a regulated power supply to drive the amp. We only need one unregulated [dual] power supply per channel in the range of +/-10 VDC to +/-20 VDC. Figure 1 in the attached PDF Headphone 006 shows the [even] simpler schematic of the transconductance amp. It is mostly like that shown in the previous thread less the now-obsolete +/- 6 VDC supply. The schematic also shows the use of one dual +/- 15VDC power supply. The schematic of this common power supply is shown in Figure 2. It is comprised of a power transformer (12Vac secondary plus a center tap), a bridge rectifier and 2 filter capacitors (4700 microfarad each). Note its the battery model which I used in Figure 1. The CENTER TAP of the transformer's secondary joins the common junction of the filter capacitors, and most importantly it is the OUTPUT PORT Vo and not the customary ground.
In operation, the variation in the output voltage Vo is a trivial disturbance to the working of the JFET constant current sources (CCSs). The +/- 200 mV peak to peak level of Vo across the headphone [for my comfortable listening] is less than 1% voltage variation across the CCSs; i.e. (0.2V signal divided by 30 V power supply times 100 = 0.7%). Also, the value of the current passing through the JFETs is quite stable over a wide range of power supplies. Thus the JFETs resist the voltage variations in Vo and present themselves to the power supply as impedances of high value.
Food for thought.
Suppose I use two resistors (5.6 K each instead of the JFET contant current sources. One resistor is connected to the +15 V and the other to -15 V. The current passing through them is 30 V divided by 11.2 K = ~2.7 mA which also flows through the diodes to bias the output transistors. In operation the situation is quite different from the one I described above.
- The 200 mV peak to peak variation in Vo is still a minor disturbance to the current flowing through the resistors (5.6 K).
- There is now a substantial AC feedback and DC servo feedback from the output Vo to the input port. Maybe beneficial!
- The input impedance of the amplifier is now 5.6 K divided by 2 = 2.8 K. It is acceptable to sources of low output impedance.
I have an omission in Figure 2 of the previous thread. Figure 2 shows the schematic of a common power supply. The secondary of the power transformer equal to 12.6 Vac plus a center tap should give ~+/-18.5 VDC instead of the quoted +/-15 VDC. I used a variac or autotransformer on the primary of the power supply transformer to tweak the output voltage to +/-15 VDC.
What value does this headphone transconductance amplifier or [attenuator; see below for more discussion] bring to the table? How does it compare with other voltage source amplifiers on several issues?
- The output port of this amp can be shorted to ground indefinitely without damage. No one dares to do this with any of the voltage source headphone amplifiers which currently populate the general heading of Headphone Systems. It is contraindicated!
- A tool to measure the Impedance versus Frequency for a loudspeaker, its individual drivers and headphones. The attached Word document ImpedanceGrado80i shows such a graph. The two headphones were connected in parallel for the measurements. The Y axis is 20 X Log of the ratio of the [voltage at any frequency] to the [voltage at 1 KHz]. The X-axis is the Log of the frequency spanning 30Hz to 20 KHz. The graph is a typical Impedance graph of an electrodynamic driver; after the sharp edges (due to measurement inaccuracies) are smoothed out. The trends are there. The driver has a peak at resonance at low frequency and a rising impedance with frequency due to the inductance of the drivers' coils. This graph may remind you of an output of a "tone control circuit". The hidden bonus in this graph is a boost in the output power by the headphone for bass and treble frequencies. By contrast; a voltage source headphone amplifier is NOT a tool to graph Impedance versus Frequency and will not claim this auto boost of bass and treble frequencies; at least for Grado 80i. The bass boost is beneficial for bass-shy headphones and the treble boost is a plus for those of us who have lost hearing in the high end due to older age. A voltage source headphone will give the user the exact opposite effect which is a simultaneous muting of the bass and treble frequencies. The tonal balance of the same headphone is different from to a voltage source and a current source devices.
- Use the circuit which I described in Headphoone001 to graph the impedance versus frequency of loudspeakers or individual loudspeaker drivers. Bypass the 220K hearing protector at its input and drive the unity gain amp with a 1-4Vpp signal. DIYers usually find the need to graph the impedance-frequency profile of woofers. This is straight forward with a sine function generator and an AC voltmeter so as to cover the important range of 20Hz to 150 Hz.
This is it for the Headphone Device.
As I made the device simpler [in design], hum from the power supply crept in the output. Hum became noticeable and/or objectionable. So, power to the device is now enabled by using rechargeable batteries; which rendered it portable if that is an important attribute to the DIYer. The attached PDF Headphone 007 shows two schematics. The first is for the active circuitry and the second details the circuit which can be used to recharge the power supply batteries when the device is idle. Let us focus on the first schematic. Note the following:
The battery-recharging circuit is straight forward. It will be useful for small 12 V Pb-acid batteries. Two commercial or home-made rechargers are needed for both channels with one recharger for each power supply rail. A [single pole triple throw] switch is used for each channel. One position recharges the batteries for both power supply rails and both audio channels. The opposite position of the switch disables both rechargers completely and simultaneously turns on power to both channels of the headphone device. It will be a useful tool to have and or build so to use instead of the classical power line supply.
The sound of the device (one channel driving both headphones), was detailed and tonally balanced. Understandably, there was no 60 Hz hum in the headphones, but a mere 2 milliVolt peak to peak broad band noise at the ouptut port as seen on the scope. I am inclined to "box up" two channels. Pictures of my progress will be forthcoming.
As I made the device simpler [in design], hum from the power supply crept in the output. Hum became noticeable and/or objectionable. So, power to the device is now enabled by using rechargeable batteries; which rendered it portable if that is an important attribute to the DIYer. The attached PDF Headphone 007 shows two schematics. The first is for the active circuitry and the second details the circuit which can be used to recharge the power supply batteries when the device is idle. Let us focus on the first schematic. Note the following:
- The TIP 120 and 125 transistors were replaced by MJE15028 (NPN) and 15029. The TIP transistors are recommended for low speed switching and may not be suitable for this application.
- The JFET constant current sources in PDF Headphone006 were replaced by resistors (4.7K). They accomplished 3 functions. The first is to supply current to the red LEDs so as to bias the output stage in Class A. the second is to provide corrective AC feedback. And thirdly is to enable a DC servo action so as to minimize DC offset at the output port Vo. The output DC offset will not rise above 100 millivolts after unplugging the headphones; this is highly protective of the device!
- Note the use of the rechargeable 9V Nickel-Metal Hydride batteries. The DIYer may wish to consider using rechargeable 12 V Lead-acid batteries instead; the gel type or other.
- Blocking capacitors in the input circuit are not recommended or needed.
- For a minimum listening level, the input impedance of the device is equal to 25 K (vol control) + 10 K + 4.7K/2 = 37K. The 10 K resistor is for my [and your] hearing protection.
- For a maximum listening level, the input impedance is ~12 K.
- My listening input impedance is variable and centered around ~[12K (vol control) + 10K + 4.7 K/2 = ~27 K]. I am using Grado SR 80i headphones. The oscilloscope trace at the output port was ~200 milliVolts peak to peak for my comfort zone. Note that the input signal Vi from the CD player resigstered upwards of 6 Volts peak to peak.
The battery-recharging circuit is straight forward. It will be useful for small 12 V Pb-acid batteries. Two commercial or home-made rechargers are needed for both channels with one recharger for each power supply rail. A [single pole triple throw] switch is used for each channel. One position recharges the batteries for both power supply rails and both audio channels. The opposite position of the switch disables both rechargers completely and simultaneously turns on power to both channels of the headphone device. It will be a useful tool to have and or build so to use instead of the classical power line supply.
The sound of the device (one channel driving both headphones), was detailed and tonally balanced. Understandably, there was no 60 Hz hum in the headphones, but a mere 2 milliVolt peak to peak broad band noise at the ouptut port as seen on the scope. I am inclined to "box up" two channels. Pictures of my progress will be forthcoming.
Above you wrote that you smoothed out the sharp edges of the impedance plot due to measurement inaccuracies. Are you sure those are measurement inaccuracies? Afaik, the impedance is not as smooth and with a transconductance amp this will cause similar "inaccuracies / sharp edges" in the frequency response.
Dunno if you have the equipment needed for this but I'd love to see a sweep that you recorded from your Grados powered by this amp.
Dunno if you have the equipment needed for this but I'd love to see a sweep that you recorded from your Grados powered by this amp.
I agree with your analysis. Your recommendation will be the ideal experiment. I have a function generator with a digital readout and an oscilloscope. I dialed the desired frequency and then measured the attendant output voltage on the scope's 0.1 V/division scale. I utilized 4 divisions each of which contains 4 subdivisions. The measurement inaccuracy was in my reading the scope's subdivisions. Was it 3.6 or 3.5 subdivisions? Please go to the website www.firstwatt.com. Mr Pass has 2 articles which discuss transconductance (current source) amps and their use to drive speakers. He showed graphs of your proposed experiment; except on high end loudspeaker drivers. Regards
Transformer-coupled headphone amp.
Please visit the forum: Pass labs, F6 Amplifier, page 42 thread #411. I inadvertently made a headphone amp in my quest to make an attenuator for a CD source. Briefly, two operational amplifiers [a dual OPA 2134] drive the primary [500 Ohm] of a transformer in an out-phase manner [schematic is attached below]. The output impedance of the transformer's secondary is 8 Ohms. I still classify this amp as a transconductance amp because of its 8 Ohm output impedance and the lack of overall loop feedback [from the secondary to the primary circuit].
If you choose to adopt this circuit:
Please visit the forum: Pass labs, F6 Amplifier, page 42 thread #411. I inadvertently made a headphone amp in my quest to make an attenuator for a CD source. Briefly, two operational amplifiers [a dual OPA 2134] drive the primary [500 Ohm] of a transformer in an out-phase manner [schematic is attached below]. The output impedance of the transformer's secondary is 8 Ohms. I still classify this amp as a transconductance amp because of its 8 Ohm output impedance and the lack of overall loop feedback [from the secondary to the primary circuit].
If you choose to adopt this circuit:
- Select a transformer with a wide frequency response [e.g. 20 Hz to 20 KHz; instead of the one I used of 300 Hz - 3.5 KHz] and a high impedance for its primary; for example 10K [instead of 500 Ohm I used ] to make it easy on the output stages of the Op Amps.
- I used point to point wiring to build 2 such prototypes [in stereo].
- One prototype has a string of 5 resistors [2 Ohms each] connected in series across the transformer's secondary output; per channel. A dual 7- position rotary switch was then used as a stepped volume control. This is the [level-controlled] output for headphone or attenuator to power amp.
Please allow me to mull [the bolded] further and answer you at a latter time with a schematic showing voltage and current values on/at the primary winding. I recall having it to do with creating a summing junction [at inverting OA] so as to satisfy zero volts [ac and dc] at the center tap of the primary as it must be for symmetry. Also I did not want to stress the output stages of both OAs by having a too low output impedance load for them. The output load for each was 500 Ohm [transformer]// 560 Ohms;~250 Ohms. Noting also my concern that the input signal to the non-inverting OA [from CD player] may be as high as 8 Vp-p!.
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