The Sorensen Audio Experiment, design description version 3.0

Contents

 

Contents. 1

1. Copyright Notice. 1

2. Introduction. 1

2.1. Operation theory. 1

3. Original requirements. 2

4. Circuit Description. 2

4.1. Overview.. 2

4.2. Error Amplifier / Integrator 3

4.3. Comparator 3

4.4. Power output stage (including MOSFET driver) 3

4.5. Output filter 3

5. Implementation Considerations. 4

5.1. Making a PCB layout…... 4

6. Other notes. 4

6.1. Tests / Measurements. 4

6.2. Design history. 5

7. Acknowledgements. 6

1.Copyright Notice

      Copyright (c)  2001-2004 Johan Sörensen.

      Permission is granted to copy, distribute and/or modify this document

      under the terms of the GNU Free Documentation License, Version 1.1

      or any later version published by the Free Software Foundation;

      with the Invariant Sections being section 1. (this copyright notice), with no

      Front-Cover Texts, and with no Back-Cover Texts.

      A copy of the license is included in the file entitled "License.htm".

2.Introduction

The aim with this design is to realize a High Fidelity, high power audio amplifier, in a simple and relatively cheap way. That is, at least cheap compared to commercial High End audio equipment. This design is the third generation in the Sorensen Audio Experiment series, therefore designated as version 3.0.

 

Note! The contents of this document describe the current state of the project, which is considered to be “finished”. There could still be a need for some improvements to this documentation, specifically in terms of how-to-build instructions. (Other suggestions are also welcome!)

Note also that I am not advocating that others should build this! You are free to do so, using the information you find here, but then you do it on your own risk!
This is not a project for a beginner!

 

2.1.Operation theory

The basic design principle is class D amplifier operation, where the output transistors alternate between only two states; completely off and completely on. This alternation is done many times per second, creating a pulse-width modulated output signal. This is then filtered, to closely reproduce the desired music signal in the audible range.

 

There are different ways of creating a pulse-width modulated signal. A common one is to compare an analog input signal with a high-frequency triangle wave, and drive the output to high when the input is higher than the momentary triangle wave voltage, and drive the output to low otherwise. This creates a pulse-width modulated output signal, with a switch frequency equal to the frequency of the triangle wave.

 

This basis for design is slightly different. The picture below shows the generalized elements of the circuit.

This circuit can be described as a comparator driving an output stage (to either “high” or “low” state), with a feedback loop around the whole thing, and a noise-shaping loop filter providing the input to the comparator. A theoretical analysis of the noise-shaping filter can be found at http://listen.to/audioexperiment.

There is nothing setting a fixed switch frequency in this picture. If all components were ideal, the switch frequency would be virtually infinite. But in reality, components are far from ideal (comparators have lag-time, transistors take time to switch on and off, and so on…). This means that all the components in the feedback loop in this design together sets the switch frequency. This is further explained below.

3.Original requirements

·         Sound quality near “High End” state of the art. (This is subjective, but usually, low THD and high damping factor is a good thing J.)

·         Power delivery capability (per channel) of at least 100W continuously in 8ohm with < 1% THD.

·         The design of the output stage, and the voltage swing increase circuit driving that, should be independent of the supply voltage. (The theoretical minimum supply voltage needed to deliver 100W in 8ohm is ± 40 Volt for a single ended output). In practice, with a conventional rectified transformer supply, a 38 V AC transformer yielding ~54 V DC after rectification is needed.

·         A well-constructed embodiment of the design should be possible to be EEC/FCC type approved.

4.Circuit Description

 

4.1.Overview

 

The descriptions in this chapter refer to the schematics diagram, version 3.0E.

 

The present design described in this document creates the pulse-width modulated output signal through self-oscillation. Oscillation is something that is normally avoided at all cost in amplifiers in general, but this case is very different.

In this design, the comparator, the voltage swing increase circuit, and the power output stage, together make a very high gain amplifier, which is (only) optimized to swing to the positive or negative supply voltage alternatively. A feedback loop is then applied around this composite amplifier, and an integrating error amplifier controls the input to the aforementioned composite amplifier. This in effect creates a very fast on-off regulation, which oscillates at nearly 1MHz in the present design. This pulse-width modulation frequency is significantly higher than the highest audible frequency.

Another way of understanding the operation of this circuit is that the error amplifier compares the integrated momentary value of the output stage with the desired value defined by the input signal, and then drives the output to the positive or negative supply respectively depending on the outcome of the comparison. Of course, driving the output to the positive supply for a while invariably and quickly leads to a integrated output value that is slightly “too high”, thus forcing the output to the negative supply instead, and so on… The point is that this process is so quick and accurate compared to the fluctuations in the input signal, that the latter is followed quite closely by the filtered output signal.

 

The remaining sections in this chapter contain some notes, explanations and/or ideas concerning the different sub-circuits of the complete design.

 

4.2.Error Amplifier / Integrator

 

The LF356 (U2) serves as an integrating error amplifier. (The integration is a critical part, in that it does noise shaping.)

The value of C8 determines the integration constant. The suggested value of C8 was determined empirically – you may try to tweak this up or down, and observe the effect on the overall amplifier operation (in particular, the audible result in terms of distortion).

 

* Note: evaluate other operational amplifiers than LF356 for this. (A component with very low voltage offset and low bias currents on the inputs is preferable here!)

 

4.3.Comparator

 

The LM311 comparator unfortunately only has an open collector output, which means that it’s output can only pull to the negative supply, and there has to be a pull-up resistor to the positive supply. To alleviate the bad positive drive capability, a BF245A JFET transistor is applied directly after the LM311 output. Since the JFET is configured as a voltage follower, it in turn has a not-so-good negative pull capability (due to R51). This is finally alleviated by D33-R47, such that the final output of this stage (the Source on J4), has a reasonably good positive as well as negative drive capability.

 

4.4.Power output stage (including MOSFET driver)

 

In this design, two complementary P- & N-channel MOSFET transistors are used at the output. These are driven by one MAX626, via a capacitive level shift and clamping circuit at the FET transistor gates.

 

The first 74HCT14 inverter, between the comparator and the two other delay matching circuits is there only as an impedance buffer.

The RC-filters, bypassed in different directions with switch diodes, preceding the two other 74HCT14 inverters, are there to introduce suitable turn-on delays for the MOSFETs. This is to avoid cross-conduction, i.e. a small dead time is allowed between the moment where one MOSFET is turned off, and the other one is turned on. The values for C19 and C20 depend on what kind of MOSFETs that are used; bigger and slower FETs will need more turn-on delay. Also, there is a notable discrepancy between the values needed for a circuit on a breadboard, and one on a PCB. Once the design has passed the beta-testing stage, I will document all component selection cross-dependencies found.

 

4.5.Output filter

 

Making an adequate output filter is as much about getting good components as anything else – in particular inductors that withstand high currents without big losses and/or distortion. I prefer the concept of core-less inductors, since there cannot be any magnetic saturation effects causing distortion (as the may be particularly in toroid inductors, depending on the core material).

Designing a 2nd order Butterworth low-pass-filter with a transition frequency of 50 kHz, yields filter component values according to the table below:

RLoad

L (uH)

C (nF)

8

36

280

4

18

560

 

I have wound my own inductor coils using 1mm thick wire (coated with a thin insulation - the kind of wire used in transformers…). 22 turns with a diameter of 4cm should yield an inductance of ~20uH.
Should you use another wire thickness, or other dimensions, the inductance can be calculated with

L = (0.08d2n2) / (3d +9l)

where

L is given in uH

d = diameter in cm

n = number of turns

l = length of the coil

5.Implementation Considerations

5.1.Making a PCB layout…

 

There are issues relating to the PCB design implementation and output filtering, which are important for the overall performance in a real embodiment of the design. Be careful how you draw the power supply and earth strips, considering where high frequency currents will flow. The abstract schematics design does not fully reflect the capacitor bypassing needed in various places for successful operation!!!

Create one ground plane for the Low Voltage parts (the ones supplied with ± 5V), and one ground plane for the High Voltage parts. Then pull wires from these two grounds to a common ground point at the power supply.

In the output filter, the ground side of C14 must be connected to the high voltage ground; this is the same ground point you use for bypassing the high voltage supplies with capacitors of at least a few tens of µF near the output transistors. Also, bypass the supplies with 330nF capacitors, with good high frequency characteristics. (As close to the output transistors as possible.)

Experience from my process of building and testing the circuit on PCBs suggests that this kind of design [i.e. switching power amp.] is quite sensitive to PCB layout (since high powers and switching currents are involved), so it may be a lengthy and arduous process test out different PCB variations to find out what is working and what is not, if you don’t know exactly what you’re doing the first time…

6.Other notes

6.1.Tests / Measurements

 

So far, only a few rudimentary tests and measurements have been made, using an oscilloscope and a normal multimeter instrument, and a PC as a signal source. (As of June 8, 2002)

The tests were made on a PCB 3.0D, with a power supply giving +/- 54V with no load (drops to 45-50V fully loaded). This was aided with a +/-30V laboratory power supply, which is further regulated down to +12V and +/-5V for the various parts of the amplifier circuit. A large 8ohm resistor served as a test load.

The output transistors were mounted on a small heat sink (approx 6 by 3 cm).

Some notes:

 

And of course, there are no THD measurements etc. yet; what I’ve done so far serves to demonstrate the quantitative (if not qualitative) abilities of the design J.

 

6.2.Design history

Just for the record, here is a summary of milestones in the development of the 3.0 design and PCBs:

Date

Event

Feb. 2001

Created the Audio Experiment Website. Started collecting ideas & background material on the making of a MOSFET switching output stage, capable of sustaining 100-200W output power into 8ohms.

Dec. 28, 2001

Started building circuit of yet-to-be 3.0A on a prototype board.

Jan. 2002

Much experimenting and discussions on the Audio Experiment forum (and offline) led to 3.0A design with MAX626 driver and capacitive coupling to gates of the output MOSFETs.

Feb. 1, 2002

Had a prototype for 3.0A on the breadboard working quite well with the limited laboratory power supply. (+/- 30V)

Feb. 11, 2002

Had a first PCB (160x100mm) with two channels ready for the 3.0A design; however, this was unstable and unreliable, and sounded bad when both channels were operated concurrently.

Feb. 24, 2002

After some testing and tweaking of the 3.0A PCB, it was found to have low efficiency (~70%) and, was generally unreliable.

Mar. 2002

Started design on a new PCB, and prepared it to use either LT1016 or LM311 as comparator. (LT1016 is much faster, but therefore also more prone to pick up noise.)

Apr. 7, 2002

Second PCB (3.0B) ready and working to some degree with the LM311 comparator.

Apr. 16, 2002

The great moment when the new design started amplifying music signals in my living room “for real”. This time quite reliably, and with (subjectively) acceptable efficiency. (The amplifier casing was only very slightly elevated in temperature even after hours of playing at high levels.)

Apr.23, 2002

However, when trying to measure efficiency, I realized that the lack of DC stability on the outputs wasn’t without problems. Thus the card had to be augmented with this feature. (I had had some foresight when designing it, so it was possible to retrofit a couple of extra transistors onto the card J.)

April 2002

Although I was quite happy with the operation of the 3.0B PCB, I started designing a new PCB (3.0C) where each channel was totally independent (on 80x100mm).

May 2, 2002

The tweaked second PCB was put into operation again, successfully.

May 25, 2002

The third PCB set had proven impossible to get to work, apparently due to PCB layout issues. (I hadn’t been as generous with ground planes as on the 3.0B PBC L.) So, a fourth design was started.

June 5, 2002

Two 3.0D PCBs were initially tested – sounding great J.

June 8, 2002

Tested operating limits and efficiency of 3.0D PCBs; see section 6.1.

Aug. 4, 2004

Clean-up of document. Removal of input buffer OP that was present in the 3.0D design.

 

7.Acknowledgements

Thanks to:

Svante Gellerstam for listening to my stuff and encouraging the project right from the start (1996!).

Daniel Ståhl for providing various electronics expertise and measurement help.

My wife for tolerating the audible activities.

My father-in-law for supporting the project,

Joseph Meisenhelder for sharing his experience on MOSFET driving techniques.